GMSK and OFDM nonlinearly and linearly amplified cellular and wireless networks

ABSTRACT

Processor, transmit filter and modulator for processing, filtering a signal into time division multiplexed (TDM) Gaussian filtered signal and into Orthogonal Frequency Division Multiplexed (OFDM) signal and for modulating Gaussian filtered signal into Gaussian Minimum Shift Keying (GMSK) modulated signal and processed OFDM signal into OFDM modulated signal. Processor for processing a signal into processed cross-correlated in-phase and quadrature-phase filtered signal and into cross-correlated in-phase and quadrature-phase spread spectrum signal. GMSK modulated signal provided to a cellular network and OFDM modulated signal provided to a different wireless network. A first amplifier and transmitter for nonlinearly amplifying and a second amplifier for linearly amplifying modulated signal and for providing signals to a transmitter. Receiver and demodulator system for receiving and demodulating transmitted GMSK modulated and OFDM modulated signal. Receiver and demodulator having two or more than two antennas and incorporating diversity reception. Receiver and demodulator system for receiving and demodulating transmitted GMSK modulated signal has a receive filter and receive processor for filtering, processing and providing demodulated cross-correlated in-phase and quadrature-phase filtered signal, Receive filter is mismatched to transmit filter.

RELATED APPLICATIONS

This Application is a Continuation Application of U.S. patentapplication Ser. No. 10/831,724, filed on Apr. 24, 2004 which is acontinuation Application of U.S. patent application Ser. No. 09/370,362,filed on Aug. 9, 1999, now U.S. Pat. No. 6,757,334 and claims thebenefit of the aforementioned applications. Related patent Applications,issued patents and other publications and references are alsoincorporated herein as listed in the Attached Supplemental InformationDisclosure (SID) and attached to the SID.

This application claims the benefit under 35 U.S.C. 119(e) of U.S.Provisional Patent Application Ser. No. 60/095,943 entitled “FQPSKTRANSCEIVERS” filed 10 Aug. 1998 [PP3]; and incorporated herein byreference.

Other related United States patent applications are co-pending U.S.patent application Ser. No. 09/111,723 [PP1] filed 8 Jul. 1998 andentitled “FMOD TRANSCEIVERS INCLUDING CONTINUOUS AND BURST OPERATEDTDMA, FDMA, SPREAD SPECTRUM CDMA, WCDMA AND CSMA”; U.S. ProvisionalPatent Application Ser. No. 60/098,612 [PP2] entitled “FK MODULATION ANDTRANSCEIVERS INCLUDING CLOCK SHAPING PROCESSORS” filed 31 Aug. 1998;each of which is hereby incorporated by reference.

In this continuation application, Applicant corrected certaintypographical errors which were noticed by Applicant in the parentpatent application.

FIELD OF THE INVENTION

This invention relates generally to Bit Rate Agile (BRA) signalprocessors; more particularly to cross-correlated signal processors forincreasing RF spectral and power efficiency of modulated transmittedsignals including but not limited to digital binary, digital multilevel,and/or analog modulated signals operated in linearized and inpower-efficient Non-Linearly Amplified (NLA) systems; and mostparticularly to BRA and RF Agile Cascaded Time Constrained Signal (TCS)response and Long Response (LR) filtered and Mis-Matched (MM) filtered(ACM) quadrature phase, frequency and amplitude modulated Transmitter,Receiver, and Transceiver systems having these characteristics andmethods and procedures provided thereby.

BACKGROUND OF THE INVENTION

The most important objectives of wireless communications, broadcasting,telemetry, infrared and in general “radio” systems as well as “wired”systems include: power and bandwidth or spectrum efficiency combinedwith robust Bit Error Rate (BER) performance in a noisy and/or stronginterference environment. These system objectives are specified innumerous systems including wireless communications and cellular systems,satellite systems, mobile and telemetry systems, broadcasting systems,cable, fiber optics and practically all communication transmissionsystems. A partial list of publications, references, and patents areprovided separately below. The cited publications, references [1-23] andpatents [P1-P8], and the references within the aforementionedpublications contain definitions and descriptions of many terms used inthis new patent disclosure and for this reason these conventional termsand definitions will be described only briefly, and highlighted on acase by case basis.

Robust or high performance Bit Error Rate (BER) specifications and/orobjectives are frequently expressed in terms of the required BER as afunction of Energy per Bit (Eb) divided by Noise Density or simply noise(No), that is, by the BER=f(Eb/No) expression. Low cost, reduced size,and compatibility and/or interoperability with other conventional orpreviously standardized systems, also known as “legacy systems,” arehighly desired. Several standardization organizations have adoptedmodulation techniques such as conventional Binary Phase Shift Keying(BPSK), Quadrature Phase Shift Keying (QPSK), Offset Quadrature PhaseShift Keying (OQPSK) also designated as Staggered Quadrature Phase ShiftKeying (SQPSK), and pi/4-QPSK (or π/4-QPSK) techniques includingdifferential encoding variations of the same. See publications [1-23]and referenced patents [P1-P8] for examples and further description. Forspectrally or spectrum efficient signaling (such as band-limitedsignaling), these conventional methods exhibit a large envelopefluctuation of the modulated signal, and thus have a large increase inpeak radiated power relative to the average radiated power. For thesereasons such systems are not suitable for BRA, robust BER performanceNLA operated RF power efficient systems.

Within the present state of the technology, for numerous BRA Transceiverapplications, it is not practical to introduce band-pass filtering afterthe NLA power efficient Radio Frequency (RF) final amplifier stage. Herewe are using the term “Radio Frequency” (RF) in its broadest sense,implying that we are dealing with a modulated signal. The RF could be,for example, as high as the frequency of infrared or fiber optictransmitters; it could be in the GHz range, for example, between 1 GHzand 300 GHz or more, or it could be in the MHz range, for example,between about 1 MHz and 999 MHz, or just in the kHz range. The term RFcould even apply to Quadrature Modulated (abbreviated “QM” or “QMOD”)Base-Band (BB) signals or to Intermediate Frequency (IF) signals.

In conventional BPSK, QPSK, OQPSK or SQPSK, and differentially-encodedphase-shift keying systems variants of these systems, such as DBPSK andDQPSK, as well as in pi/4-DQPSK and trellis coded QPSK and DQPSK, largeenvelope fluctuations require linearized (LIN) or highly lineartransmitters including frequency up-converters and RF power amplifiersand may require expensive linear receivers having linear Automatic GainControl (AGC) circuits. A transmitter NLA reduces the time domainenvelope fluctuation of conventional QPSK type of band-limited signalsand this reduction of the envelope fluctuation, being a signaldistortion, is the cause of spectral restoration or spectral regrowthand the cause of unacceptably high levels of out-of-band spectral energytransmission, also known as out-of-band interference. Additionally, forconventional BPSK, QPSK, and also Quadrature Amplitude Modulation number(QAM) signals, undesired in-phase channel (I) to quadrature channel (Q)crosstalk is generated. This crosstalk degrades the BER=f(E_(b)/N₀)performance of the modulated radio transmitter.

Experimental work, computer simulation, and theory documented in manyrecent publications indicates that for band-limited and standardizedBPSK, QPSK, OQPSK or SQPSK or pi/4-QPSK, and QAM system specifications,very linear amplifiers are required to avoid the pitfalls of spectralrestoration and of BER degradation. Linearized or linear amplifiers areless RF power efficient (during the power “on” state, power efficiencybeing defined as the transmit RF power divided by DC power), areconsiderably more expensive and/or having less transmit RF powercapability, are larger in size, and are not as readily available as NLAamplifiers. The advantages of NLA over LIN amplifiers are even moredramatic at higher RF frequencies, such as frequencies above about 1 GHzfor applications requiring low dc voltage, for example applications orsystems operating on size “AA” batteries having only 1.5 Volt dc and forhigh RF modulated power requirements, for example transmit RF power inthe 0.5 Watt to 100 Watt range.

Published references [P1 to P8] and [1 to 23] include additionalbackground information. These references include descriptions ofbinary-state and multiple-state Transmitter/Receiver (Transceiver) orfor short (“TR”) systems that are suitable for NLA. In theaforementioned references Processors, Modems, Transmitters, Receiversand Transceivers, suitable for NLA, have been described, defined anddesignated as first generation of Feher patented Quadrature Shift Keying(FQPSK). For example, in reference [22] published on May 15, 1999 theauthors Drs. M. K. Simon and T. Y. Yan of JPL/NASA-Caltech present adetailed study of Unfiltered Feher-Patented Quadrature Phase ShiftKeying (FQPSK). In references [1-22] and patents #[P1-P8] numerous firstgeneration FQPSK technology based terms, and terms other than the FQPSKabbreviation/acronym have been used. In addition to FQPSK Transceivers,these first generation of systems have been also described and/ordefined as: Feher's Minimum Shift Keying (FMSK), Feher's Frequency ShiftKeying (FFSK), Feher's Gaussian Minimum Shift Keying (FGMSK), Feher'sQuadrature Amplitude Modulation (FQAM) and/or Feher's (F)Modulation/Amplification (FMOD). Additionally terms such as SuperposedQuadrature Amplitude Modulation (SQAM), Intersymbol Interference andJitter Free (IJF) and/or IJF-OQPSK have been also described in Feher etal.'s prior patents and publications, each of which is incorporated byreference.

In the cited patents and other references, among the aforementionedabbreviations, acronyms, designation, terms and descriptions the “FQPSK”abbreviation/term has been most frequently used to describe in mostgeneric terms one or more of these afore described Feher or Feher et al.first generation of Non-Linearly Amplified (NLA) inventions andtechnologies. The 1_(st) generation of FQPSK systems have significantlyincreased spectral efficiency and enhanced end-to-end performance ascompared to other NLA systems. RF power advantages, robust BERperformance, and NLA narrow spectrum without the pitfalls ofconventional BPSK and DBPSK, QPSK and OQPSK have been attained withthese 1^(st) generation FQPSK systems and methods. The generic 1^(st)generation terms such as FQPSK, as well as other previously mentionedterms/acronyms are retained and used in this description to describe thenew BRA, Code Selectable (CS), Modem Format Selectable (MFS) andmodulation-demodulation Mis-Matched (MM) filtered architectures andembodiments of “2^(nd) generation” FQPSK Transceivers.

While these earlier issued patents and publications describe material ofa background nature, they do not disclose the original new enhancedperformance bit rate agile and modulation agile/selectable technologiesdisclosed in this new invention.

Partial List of Relevant Literature

Several references, including United States patents, International orForeign Patents, publications, conferences proceedings, and otherreferences are identified herein to assist the reader in understandingthe context in which the invention is made, some of the distinctions ofthe inventive structures and methods over that which was known prior tothe invention, and advantages over the invention. No representation ismade as to whether the contents of the cited references representprior-art as several of the cited references have a date after theeffective filing date (priority date) of this patent application. Thislist is intended to be illustrative rather than exhaustive.

United States Patents

[P1] U.S. Pat. No. 5,784,402 Issued July 1998 to Feher

[P2] U.S. Pat. No. 5,491,457 Issued February 1996 to Feher

[P3] U.S. Pat. No. 4,720,839 Issued January 1988 to Feher et al.

[P4] U.S. Pat. No. 4,644,565 Issued February 1987 to Seo/Feher

[P5] U.S. Pat. No. 4,567,602 Issued January 1986 to Kato/Feher

[P6] U.S. Pat. No. 4,350,379 Issued September 1982 to Feher

[P7] U.S. Pat. No. 4,339,724 Issued July 1982 to Feher

[P8] U.S. Pat. No. 3,954,926 Issued March 1976 to Feher

Foreign Patent Documents

[PF1] Canadian Patent No. 1130871 August 1982 [PF2] Canadian Patent No.1211517 September 1986

[PF3] Canadian Patent No. 1265851 February 1990

Other Publications

-   1. Feher, K.: Wireless Digital Communications: Modulation Spread    Spectrum. Prentice Hall, 1995.-   2. Feher, K.: Digital Communications: Satellite/Earth Station    Engineering. Prentice Hall, 1983. Available from Crestone    Engineering-Noble Publishing, 2245 Dillard Street, Tucker, Ga.    30084.-   3. Feher, K.: Advanced Digital Communications: Systems and Signal    Processing. Prentice Hall, 1987. Available from Crestone    Engineering-Noble Publishing, 2245 Dillard Street, Tucker, Ga.    30084.-   4. Feher, K.: Digital Communications: Microwave Applications.    Prentice Hall 1981. Since 1997 available from Crestone    Engineering-Noble Publishing, 2245 Dillard Street, Tucker, Ga.    30084.-   5. Feher, K. and Engineers of Hewlett-Packard: Telecommunications    Measurements, Analysis, and Instrumentation: Prentice Hall 1987.    Since 1997 reprints have been available from Crestone    Engineering-Noble Publishing, 2245 Dillard Street, Tucker, Ga.    30084.-   6. Feher, K., Emmenegger, H.: “FQPSK Use for Electronic News    Gathering (ENG), Telemetry and Broadcasting,” Proc. of the National    Association of Broadcasters NAB'99 Broadcast Engineering Conference,    Las Vegas, Apr. 19-22, 1999.-   7. Feher, K.: “FQPSK Doubles Spectral Efficiency of Operational    Systems: Advances, Applications, Laboratory and Initial    Air-to-Ground Flight Tests” (Date of Submission: Aug. 14, 1998).    Proc. of the International Telemetry Conference, ITC-98 ITC/USA 98,    San Diego, Calif., Oct. 26-29, 1998.-   8. W. Gao, S. H. Wang, K. Feher: “Blind Equalization for FQPSK and    FQAM Systems in Multipath Frequency Selective Fading Channels,”    Proc. Internat. Telemetry Conf. ITC/USA '99, Oct. 25-28, 1999, Las    Vegas, Nev.-   9. Terziev, G., Feher, K.: “Adaptive Fast Blind Feher Equalizers    (FE) for FQPS K,” Proc. Of the International Telemetry Conference    ITC USA '99, Oct. 25-28, 1999, Las Vegas, Nev.-   10. Feher, K.: “FQPSK Transceivers Double the Spectral Efficiency of    Wireless and Telemetry Systems” Applied Microwave & Wireless    Journal, June 1998.-   11. Seo, J-S. and K. Feher: “Bandwidth Compressive 16-State SQAM    Modems through Saturated Amplifiers,” IEEE Radio Commun., ICC '86,    Toronto, June 1986.-   12. Kato, S. and K. Feher: “XPSK: A new cross-correlated PSK,” IEEE    Trans. Com., May 1983.-   13. Law, E. L., U.S. Navy: “Robust Bandwidth Efficient Modulation”    European Telemetry Conference, ETC-98, Germany, May 1998.-   14. Feher, K.: “FQPSK Doubles the Spectral Efficiency. of    Operational Telemetry Systems,” European Telemetry Conference,    ETC-98, May 1998, Germany.-   15. Do, G. and K. Feher: “FQPSK-GMSK: Wireless System Tests an ACI    Environment,” Proc. of Wireless Symposium, Santa Clara, Calif., Feb.    9-13, 1998.

16. Law, E. and K. Feher: “FQPSK versus PCM/FM for AeronauticalTelemetry Applications: Spectral Occupancy and Bit Error ProbabilityComparisons” Proc. of ITC-97, Las Vegas, October 1997.

-   17. Feher, K “FQPSK Doubles Spectral Efficiency of Telemetry:    Advances and Initial Air to Ground Flight Tests,” ITC USA 98, Proc.    of the Internat. Telemetry Conference, San Diego, October 1998.-   18. Law, E. and K. Feher: “FQPSK versus PCM/FM for Aeronautical    Telemetry Applications; Spectral Occupancy and Bit Error Probability    Comparisons,” Proc. of the Internat. Telemetry Conf., Las Vegas,    Nev., Oct. 27-30, 1997.-   19. Martin, W. L., T-Y. Yan, L. V. Lam: “Efficient Modulation Study    at NASA JPL,” Proc. of the Tracking, Telemetry & Command Systems    Conference, European Space Agency (ESA), June 1998.-   20. Law, E. L., ITC-98 Session Chair: “RCC Alternate Standards and    IRIG106 update,” Briefings by DoD during ITC USA 98 Internat.    Telemetry Conference, San Diego, October 1998.-   21. K. Feher: “FQPSK Doubles Spectral Efficiency of Operational    Systems: Advances, Applications, Laboratory and Initial Air to    Ground Flight Tests”, File: ITC.98.Final Paper. Rev. 5. Aug. 14,    1998 (Date of Submission) for publication in the Proc. of the    International Telemetering Conference, ITC-98; San Diego, Oct.    26-29, 1998-   22. Simon, M. K, Yan, T. Y. “Performance Evaluation and    Interpretation of Unfiltered Feher-Patented Quadrature Phase-Shift    Keying (FQPSK),” California Institute of Technology, JPL-NASA    publication, TMD Progress Report 42-137, Pasadena, Calif., May 15,    1999.-   23. Winters, J. H.: “Adaptive Antenna Arrays for Wireless Systems,”    Tutorial Notes presented/distributed at the 1999 IEEE Vehicular    Technology Conference, Houston, Tex., May 16, 1999.

SUMMARY OF THE INVENTION

This invention includes disclosure of new and/original spectralefficient and RF power efficient-high performance technologies, newarchitectures, embodiments and new Bit Rate Agile (BRA) implementationof 2^(nd) generation FQPSK Transceivers. These inventive structures,methods, and technologies are suitable for a large class ofimplementations and applications. Numerous embodiments of the inventivestructures and methods are enabled. These include cost effectivesolutions for BRA, Modulation and Demodulation (Modem) Format Selectable(MFS) and Coding Selectable (CS) processors, modulators/demodulators,Transceivers, having agile/tunable RF frequency embodiments and aresuitable for power efficient NLA systems.

The terms abbreviations and descriptions used in the 1^(st) generationof Feher et. al inventions, highlighted in the “Background of theInvention” section, as well as other previously identifiedterms/acronyms and abbreviation and in particular FQPSK and relatedterms, used in the cited references, are retained and/or slightlymodified and are used relative to this disclosure of the new inventionto describe second generation “2^(nd) generation” BRA architectures andembodiments of FQPSK, FGMSK and FQAM Transceivers. This disclosurecontains embodiments for further significant spectral savings andperformance enhancements, and new functions and architectures, whichwere not, included in the referenced prior art patents, inventions andpublications.

BRA or “Bit Rate Agile” abbreviation and term describes technologies,implementations, embodiments suitable for design, use and applicationsin which the information rate, source rate, or the often usedalternative terms “bit rate”, “symbol rate”, or “data rate” may beselectable or programmable by the user or by one or more controlsignal(s). Bit Rate Agile (BRA) systems may be programmable by softwareor have predetermined or “selectable,” i.e., “agile” bit rateapplications. The term “bit rate agility” refers to variable and/orflexible selectable bit rates (again bit rate, symbol rate, data rate,information rate, source rate, or equivalent); the bit rates could beselected on a continuous fashion in small increments and/or in steps.These systems are designated as BRA (or Bit Rate Agile) systems. BRA,MFS, and CS systems requirements are increasing at a rapid rate.

Changeable (or variable, or selectable) amounts of cross-correlationbetween Time Constrained Signal (TCS) response processors and/orcombined TCS and Long Response (LR) processor and/or post processorfilters of in-phase (I) and quadrature (Q) phase signals of BRATransceivers, MFS and CS baseband signal processing implementations andarchitectures for tunable RF frequency embodiments having enhancedspectral efficiency and end-to-end performance are disclosed. These newBRA, MFS and CS classes of FQPSK signal processors, modems andtransceivers, with Adaptive Antenna Arrays (AAA) and RF power efficientamplifiers and entire Transceivers, operated in fully saturated or NLAmode, with intentionally Mis-Matched (MM) modulation and demodulationfilters, transmit BRA and receive BRA filters/processors, disclosedherein, attain high performance advantages and significant spectralsavings.

A changeable amount of cross-correlation between the BRA and MFS TimeConstrained Signal (TCS) response processor and/or combined TCS and LongResponse (LR) processor and/or post processor filters of the transmitterwith selectable MM between the BRA transmitter and BRA receiver and CSprocessors, including single and separate in-phase (I) and quadrature(Q) signal storage/readout generators and single and/or separate I and Qchannel D/A architectures and a bank of switchable filters forcross-correlated BRA, MFS and CS formats are also disclosed.

These new classes of 2^(nd) generation of FQPSK signal processors,modems and transceivers, with Adaptive Antenna Arrays (AAA) and RF powerefficient amplifiers and entire Transceivers, operated in BRA, MFS andCS fully saturated NLA mode, or with LIN mode with intentionallyMiss-Matched (MM) transmit BRA and receive BRA filters/processors,disclosed herein, have robust performance and significant spectralsaving advantages.

In addition to digital embodiments, BRA analog cross-correlationimplementations and combined digital-analog active and passiveprocessors, for 2^(nd) generation FQPSK Transceivers are also disclosed.Subsets, within the generic 2^(nd) generation of the FQPSK family ofprocessors, modems and transceivers are also designated as 2^(nd)generation BRA Feher's Minimum Shift Keying (FMSK), Feher's GaussianMinimum Shift Keying (FGMSK), Feher's Frequency Shift Keying (FFSK) andFeher's Quadrature Amplitude Modulation (FQAM).

Switched BRA, selectable Cross-Correlation (CC or Xcor) transmit andreceive bandwidth Mis-Matched (MM) low-pass, band-pass and adaptivefilter means and controller circuits and algorithms for preamblecontained and differentially encoded and/or Forward Error Correction(FEC) with Redundant and Pseudo-Error (PE) based Non Redundant Detection(NED) implementations for FQPSK are also described.

The term “Mis-Matched” (MM) designates an intentional and substantialmis-match (MM) between the bandwidth and/or frequency or phase responseof modulator filters and demodulator filters and/or mis-match (MM)between one or more implemented FQPSK filter(s) and the theoreticaloptimal performance minimum bandwidth Nyquist filters.

The term “Agile Cascaded Mis-Matched” (ACM) designates the BRA and RFagile (“flexible” or “tunable” RF frequency) cascaded TCS response andLR processor/filter(s) which are mismatched within their respectiveapplication and/or use within this invention.

For NLA and for LIN amplifiers, selectable FQPSK filtering strategies inthe transmitter and separately in the receiver lead to further spectralefficiency enhancements. Fast synchronization systems and robustefficient adaptive equalizers/adaptive switched systems are alsodisclosed.

The inventive structure and method includes transmit elements, receiveelements, and transmit and receive elements, and may be applied to avariety of communication applications, including, but not limited to,wireless communications and cellular systems, satellite systems, mobileand telemetry systems, broadcasting systems, cable system, fiber opticsystems, and more generally to nearly all communication transmissionand/or receiving systems.

In one embodiment of the invention, a bit rate agile communicationsystem is provided and includes a splitter receiving an input signal andsplitting the input signal into a plurality of baseband signal streams,and a baseband signal processing network receiving the plurality ofbaseband signal streams and generating cross-correlated cascadedprocessed and filtered bit rate agile (BRA) in-phase andquadrature-phase baseband signals. In another embodiment, a quadraturemodulator receiving and quadrature modulating the cross-correlatedfiltered in-phase and quadrature-phase baseband signals to generate aquadrature modulated output signal is also provided. In anotherembodiment, the baseband signal processing network includes across-correlator and at least one bit rate agile cascaded mis-matched(ACM) modulator filter.

In yet another embodiment, the invention provides a bit rate agilecommunication system including a baseband signal processing networkreceiving parallel baseband signal streams and generating combined TimeConstrained Signal (TCS) response and Long Response (LR) filteredin-phase and quadrature-phase baseband signals. In a variation of thisembodiment, the inventive structure also includes a quadrature modulatorreceiving and quadrature modulating the Time Constrained Signal (TCS)response and Long Response (LR) filtered in-phase and quadrature-phasebaseband signals to generate a quadrature modulated bit rate agileoutput signal.

In still another aspect, the invention provides a method for generatingbit rate agile signals in a communication system. The method includesthe steps of processing a plurality of signal streams to generatecross-correlated signals having changeable amounts of filtering for bitrate agile in-phase and quadrature-phase baseband signals. The inventivemethod may also include the step of receiving an input signal andconverting the input signal into the plurality of signal streams. It mayalso optionally include the further step of modulating thecross-correlated filtered in-phase and quadrature-phase baseband signalsto generate a quadrature modulated bit rate agile output signal.

In yet another aspect, the invention provides a method for generatingbit rate agile signals in a signal transmission system, where the methodincludes the steps of receiving a plurality of signal streams,processing the plurality of signal streams to generate cascaded TimeConstrained Signal (TCS) response and Long Response (LR) filteredin-phase and quadrature-phase baseband signals; and modulating the TimeConstrained Signal (TCS) response and Long Response (LR) filteredin-phase and quadrature-phase baseband signals to generate a quadraturemodulated bit rate agile output signal.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 a is a diagram depicting an Agile Cascaded Mis-Matched (ACM)enhanced spectral efficiency, high performance Transmitter/Receiver(Transceiver) block diagram for a generic class of modulated systems.

FIG. 1 b illustrates a somewhat generic Transmitter (Tx)block-implementation diagram of the invention for bit rate agile,selectable modulation formats, for hardware, firmware and/or softwareimplementations including optional single and multi-tone inserts at oneor more locations.

FIG. 2 depicts a Bit Rate Agile (BRA) integrated Base-Band Processor(BBP) with BRA post filters and BRA quadrature modulators. Signalsplitting and serial/parallel BRA converters and TCS response processorsin cascade with LR filter processors, which are MM enhancements andalternatives to referenced patents [P1 and P2] are also included.

FIG. 3 shows TCS response and cascaded LR filter baseband processor andfilter preceded by a cross-correlator, and multiplexers for I and Qsignal generation.

FIG. 4 illustrates coded baseband processing of FQPSK, FQAM, FGMSK andFMSK signals including non-redundant trellis coding of the basebandprocessed filtered I and Q signals.

FIG. 5 Time Constrained Signal (TCS) patterns, based on referencedpatents [P4; P5; P7] are illustrated.

FIG. 6 shows an Agile Cascaded Mis-Matched (ACM) implementation blockdiagram embodiment of this invention with cascaded, switched transmit(Tx) and receive (Rx) Low-Pass-Filters (LPF) for BRA applications inconjunction with Cross-Correlated and other non-cross-correlatedcascaded Time Constrained Signal (TCS) response processors and LongResponse (LR) filters having substantially Mis-Matched (MM) modulationand demodulation filters.

FIG. 7 is an alternate embodiment-block diagram of the currentinvention, including Cross-Correlator in cascade with a 2^(nd) processorand with Digital to Analog (D/A) converters followed by spectral andpulse shaping bit rate agile LR filters, implemented by Low-Pass-Filters(LPF) in the I and Q channels of this Quadrature modulator.

FIG. 8 shows the implementation of an alternate Quadrature Modulator(QM) Transceiver having one or more Intersymbol-Interference and JitterFree (IJF), Superposed Quadrature Amplitude Modulation (SQAM) or TCSresponse cascaded with LR response filter embodiments in the I and Qtransmit Baseband Processor (BBP).

FIG. 9 illustrates a Non Return to Zero (NRZ) signal pattern, a TCSresponse pattern, a signal pattern of a TCS filtered by a conventionalLow-Pass-Filter (LPF) resulting in cascaded TCS response and LongResponse (LR) pulse patterns with some overshoots. In the non-Xcor caseas well as in the Cross-Correlated case a Peak Limiter (PL) or gradualSoft Limiter or Xcor Soft Limiter reduces the amplitude peaks. The LRfilter extends the TCS response to multiple pulses and may introduceadditional spectral saving, however could introduce additional ISI andincreased peak variations.

FIG. 10 a shows a Cross-Correlated (Xcor) embodiment with Gaussian LPFand Integrator as well as sin and cos look-up tables. The TCS outputs ofthe I and Q baseband signals are further spectrally shaped and limitedby sets of I and Q channel filters. A single or multiplecross-correlator (X), cos and/or sin inverter (XCSI) is used in thisAgile Cascaded Mis-Matched (MM) (ACM) implementation.

FIG. l0 b shows the implementation architecture of an alternative XCSIwith PL single or multiple XCSI and/or Peak Limiter (PL) processor.

FIG. 11 a illustrates NRZ signal and shaped Feher Return-to-zero (FRZ)signal patterns and an embodiment having TCS response and cascade LRfilters in which the LR filter is implemented with digital IIR and/orFIR filters.

FIG. 11 b shows a pre-processor based architecture of a BRA system withsingle or multiple Signal Element (SE) storage and/or inverter and ofsingle or multiple D/A based architecture having selectable 1 to Nfilter channel banks.

FIG. 11 c shows an implementation of a Bit Rate Agile (BRA)pre-processor with single or multiple wavelets Signal Element (SE)storage and/or inverter and of filtered SE and ACM processors.

FIG. 12 a analog implementation components for cross-correlated and/orTCS-filtered data patterns and signals for bit rate agile and for highbit rate applications are shown.

FIG. 12 b shows an analog BRA baseband implementation alternative of aTCS response processor for cross-correlated or not cross-correlated Iand Q signals with selection or combined cascaded LR filter embodimentof this invention.

FIG. 13 a is a mixed analog and digital circuit implementationalternative of this cross-correlated TCS response processor in cascadewith LR filters.

FIG. 13 b shows an other alternative implemented cross-correlated systemwith a combination and/or selection of analog and digital circuits.

FIG. 13 c shows a detailed implementation structure for one of thechannels (I or Q) of a TCS response processor in cascade with a LRfilter. In some embodiments the TCS processor contains cross-correlationbetween the I and Q channels while in other implementations there is nocross-correlation between the TCS response and cascaded LR filteredsignals.

FIG. 14 shows a BRA implementation alternative with a TCS processor, oneor more D/A devices in cascade with a bank of switchable LRfilters/processors and switchable Linearized Phase or Phase Linear (PL)and Not Linear Phase (NLP) Filters.

FIG. 15 is a detailed implementation diagram alternative of this ACMmode architecture having cross-correlated TCS response waveletgenerators in cascade with LR filters.

FIG. 16 shows four (4) illustrative signaling elements generated byanalog TCS response cross-correlators, prior to the cascaded LR filters.The shown signaling elements or wavelets are for FQPSK signalgeneration, having a cross-correlation parameter of A=0.7.

FIG. 17 shows four (4) cross-correlated BRA signaling elements of one ofthe TCS response analog generated circuits for enhanced performanceFGMSK. Only 4 signaling elements are required in this BRA reducedspectrum Feher Gaussian Minimum Shift Keying (FGMSK) signal generation,having a BTb=0.5 parameter.

FIG. 18 shows Differential Encoding (DE) and Differential Decoding (DD)for FQPSK and FGMSK.

FIG. 19 eye diagrams of DE prototype BRA transmit signals for FGMSK withBTb=0.3, and FQPSK with a cross-correlation parameter A=0.7, prior toadditional baseband processing and prior to the baseband LR filter ofthe transmitter are shown. The eye diagram at the TCS response processoroutput and the I and Q cross-correlated eye diagrams, at the outputs ofthe cascaded TCS response and LR filters of FQPSK-B systems, operated inACM mode, as well as the corresponding vector constellation diagrams arealso shown.

FIG. 20 shows an FQAM implementation architecture diagram formulti-state Cross-Correlated FQPSK transmitters, also designated asFQAM. In this embodiment a single RF Amplifier operated in fullysaturated NLA mode is used.

FIG. 21 is an alternate implementation diagram of multi-state FQPSK,FGMSK and FQAM transmitters. In this embodiment two or more RFamplifiers operated in NLA saturated and/or in partly linearized (LIN)mode of operation are used.

FIG. 22 is an “Over the Air Combined” implementation architecture ofFQAM signal generation having two or more quadrature FQPSK and/or FGMSKmodulators, two or more RF amplifiers and two or more transmit antennas.

FIG. 23 shows the embodiment of an Orthogonal Frequency DivisionMultiplexed (OFDM) embodiment with FDM signal combining of a number ofFQPSK signals. In one of the embodiments of this invention RF Combiningis implemented by hardware RF components while in an alternativeimplementation the RF combining is implemented “over the air”.

FIG. 24 shows a transmit Antenna Array and/or RF Combiningimplementation of multiple modulated signals. This figure illustratesmultiple TCS response and/or cascaded TCS response and LR filteredcross-correlated baseband signal processors connected to an antennaarray and/or RF combiner.

FIG. 25 shows an ACM and PL architecture and embodiment for trelliscoded filtered cross-correlated I and Q baseband signal generation,containing TCS response and cascaded LR filters, for FQPSK, FGMSK aswell as FQAM signal generation.

FIG. 26 shows the Power Spectral Density (PSD) of NLA illustrative datalinks operated at 13 Mb/s rate per link, in the US Government authorizedband between 2200 MHz to 2290 MHz. With telemetry standardized filteredPCM/FM 3 links can be used simultaneously, with FQPSK-B the number oflinks is doubled to 6, while with NLA 16-state FQAM, also designated asa 2^(nd) generation multi-state FQPSK or FQPSK.2.4, the number of 13Mb/s links is quadrupled to 12 links (over that of standardized PCM/FM).

FIG. 27 The PSD and Integrated Adjacent Channel Interference (ACI) ofhardware measured prototype FQPSK-B in a NLA transmitter and of a BRAlinearized transmit FQPSK is illustrated in the upper figure. In thelower part of this figure the Integrated ACI of FQPSK systems with thatof GMSK (BTb=0.25) systems is compared. For NLA transmitters, theresults show a very significant (approximately 2 to 1) RF spectralefficiency advantage of FQPSK over that of GMSK systems. Theaforementioned FQPSK 2 to 1 spectral advantage over that of conventionalGMSK is measured for a typical −60 dB specification of the ACI.

FIG. 28 Spectral results of 16 state NLA systems are illustrated. TheACI results FQAM, obtained after fully saturated NLA, are compared withthat of NLA conventional pre-modulation filtered 16-state QAM systems.The spectral efficiency advantage of the NLA illustrated FQAM is morethan 200% over that of NLA prior QAM.

FIG. 29 shows BER performance curves, in terms of the customaryBER=f(Eb/No) performance curves, of FQPSK Transceivers. Hardwaremeasurements and/or computer design/software generated data andtheoretical study results show that NLA practical RF hardware FQPSKTransceivers, with intentionally and substantially Mis-Matched(MM)filters are within about 0.5 dB to 1 dB of the ideal theoretical LINamplified QPSK systems.

FIG. 30 shows a demodulation architecture for FQPSK, FGMSK and FQAM andfor other signals.

FIG. 31 shows an alternate A/D converter based demodulator architecture.This implementation/embodiment is suitable for “software radio”demodulation and/or for firmware or hardware, or combined hybridimplementations of this invention.

FIG. 32 shows transmit Antenna Arrays (AA) and receive Adaptive AntennaArrays (AAA) in this multiple transmit and receive omni-directionaland/or sectorized or high gain directional antenna-embodiment of thisinvention. This architecture has the potential to increases the NLAspectral efficiency of FQPSK, FGMSK and FQAM systems to more than 30b/s/Hz.

FIG. 33 shows a Pseudo-Error (PE)-Non-Redundant Error Detection (NRED)circuit embodiment for on-line or in-service monitor, for PE basedadaptive equalization control and for diversity control unitimplementations.

FIG. 34 shows an adaptive equalizer circuit embodiment of thisinvention. The adaptive equalizer, designated as Feher Equalizer (FE)generates the control signals in a PE based NRED circuit and is suitablefor fast adaptive equalization.

FIG. 35 is a switchable delay based embodiment of a combined adaptiveequalizer/adaptively selectable switched receiver designated as FeherRake “FR.” A PE based NRED or other NRED circuits are used forgenerating the control and switch selection signals.

FIG. 36 shows an implementation architecture for multiple adaptive FEand FR circuit embodiments with multiple demodulators.

FIG. 37 is a block diagram implementation of a two branch diversityreceiver with an adaptive equalizer and a single demodulator. A NREDbased circuit generates “smart” diversity selection and/or controlsignals.

DETAILED DESCRIPTION OF EMBODIMENTS OF THE INVENTION IntroductoryDescription of Embodiments of the Invention

This invention relates in part to Bit Rate Agile (BRA) signal processorsand particularly to cross-correlated (abbreviated “CC” or “Xcor”) and toAgile Cascaded Mis-Matched (ACM) signal processors for increasing the RFspectral efficiency and RF power efficiency of modulated transmittedsignals including digital binary, and digital multilevel signals, and ofanalog modulated signals operated in linearized (LIN) and in powerefficient Non-Linearly Amplified (NLA) systems. Cross-correlatedquadrature phase, frequency, and amplitude modulated Transmitter andReceiver (Transceiver) systems are described. The use of sectionheadings is for convenience only and is not intended to delimit thediscussion of particular aspects of the invention as aspects, features,embodiments, advantages, and applications are described through thespecification and drawings. Acronyms are used extensively throughout thedescription to avoid excessively long descriptive phrases. In someinstances the same acronym is used for a system component or structureas well as as a adjective or other qualifier for the function oroperation performed by that structure. The meaning will be clear fromthe context of the description.

The disclosed BRA systems, designated as belonging to the class ofFeher's Quadrature Phase Shift Keying (FQPSK) systems, and also by otheracronyms and abbreviations, described herein, include BRA TimeConstrained Signal (TCS) response processors and cascaded TCS and BRALong Response (LR) processors and/or post-processor filters of in-phase(I) and of quadrature (Q) phase signals of BRA transmitters andreceivers (or Transceivers when transmitter and receiver are combined).Modulation and demodulation (Modem) Format Selectable (MFS) and CodingSelectable (CS) baseband signal processing implementations andarchitectures for tunable RF frequency embodiments having Pseudo-Error(PE) based Non-Redundant Error Detection (NRED) implementationstructures are also disclosed. BRA demodulation filters Mis-Matched (MM)to that of the modulator filters, filters MM to that of theoreticaloptimal minimum bandwidth Nyquist filters. PE controlled adaptiveequalizers and diversity systems having enhanced spectral efficiency andend-to-end performance are also described and included within the scopeof this invention.

In general terms the present invention discloses and provides structureand method for cost effective solutions for Bit Rate Agile (BRA),modulation and demodulation (Modem) Format Selectable (MFS) and CodingSelectable (CS) processors, modulators/demodulators (modems),transmitters and receivers (Transceivers) with agile cascaded mismatched(ACM) architectures having agile/tunable RF frequency embodiments andwhich are suitable for power efficient systems. This disclosure containsnumerous structural and methodological embodiments and implementationarchitectures which lead to: (i) significant RF spectral savings, (ii)performance enhancements, and (iii) new features, functions andarchitectures; none of which were suggested or disclosed in the citedissued patents, inventions and references. Several of the referenceshave been described as 1^(st) generation of Feher's Quadrature PhaseShift Keying (FQPSK), Feher's Quadrature Amplitude Modulation #(FQAM),Feher's Gaussian Minimum Shift Keying (FGMSK), and Feher's Minimum ShiftKeying (FMSK) Transceivers.

The new implementation architectures, embodiments and new BRAtechnologies described in this disclosure are designated as subsets ofsecond generation of FQPSK systems, suitable for BRA operation.

Overview of Exemplary Embodiments Described

A detailed disclosure of implementation architectures and embodiments ofthis invention is contained in the following sections. In many instancesthe text is related to the description of the respective figures and tothe implementation of ACM transceivers. A changeable amount ofcross-correlation between the BRA, CS, MFS and TCS response processorand/or combined or cascaded TCS and LR filters and/or post processorfilters of the transmitter with selectable MM between the BRAtransmitter and BRA receiver and CS processors, including single andseparate in-phase (I) and quadrature (Q) signal storage/readoutgenerators and single and/or separate I channel and Q channel D/Aarchitectures for cross-correlated BRA, MFS and CS formats are alsodescribed. In agile cascaded mismatched (ACM) designs for NLA and forLinearized (LIN) amplifiers selectable FQPSK filtering strategies, inthe transmitter and separately in the receiver lead to furtherimprovements in spectral efficiency.

Within an interface unit (IU) 107 in FIG. 1 a a generalized blockdiagram of an embodiment of this inventive Transceiver 100 is shown andincludes a BRA 102, MFS, CS, MM filtered and RF frequency agile enhancedspectral efficiency, high performance Transceiver block diagram. In thisembodiment one or more input signals 150-1, 150-2, . . . , 150-N arereceived on single or multiple lead(s) 101-1, 101-2, . . . 101-N andprovided to a transmit section 152 of Interface Unit (IU) 102. The termlead or leads generally refers to a coupling or connection between theelements, and may for example refer to a wire, integrated circuit trace,printed circuit trace, or other signal link, connection, or couplingstructure or means as are known in the art. Note that the IU provides anInterface port for transmission, designated as 102 and in the Receivesection, also designated as 102, it contains the required IU receiveprocessors and provides an output interface port. The input and outputleads may contain and communicate analog or digitized voice, music,data, video, telemetry or other signals, or combinations thereof. In theFIG. 1 a embodiment, signals 150 on input leads 101 may also representSpread Spectrum, CDMA, WCDMA, CSMA, TDMA, FDMA or continuous singlechannel “Clear Mode”, or other signals such as FDM and orthogonallyfrequency division multiplexed Orthogonally Frequency DivisionMultiplexed (OFDM) signals and IU port configurations.

The BRA 102 unit, shown in FIG. 1 a provides signals 153 to a Base-BandProcessor (BBP) unit 103. This unit 103, receives in addition to theoutput signals from the IU 102, signals on lead 104 a Clock (C), on lead105 one or more Control (CTL) signals and on lead 106 one or moreSampling (SAM) signals. The combination of the aforementioned C, CTL andSAM signals is also designated with a common further abbreviation as“C”.

BBP unit 103 provides a new class of BRA Cross-Correlated (CC) signals,including ACM filtered signals. The BBP provides signals to theQuadrature Modulator (QM) unit 109. Numerous embodiments of QM 109 havebeen described in the prior art and/or in the listed references. The QMimplementation in baseband, IF and RF frequency ranges is well known bymeans of analog, digital and combined analog or digital techniques inhardware, firmware and in software and does not require furtherdescription. The Frequency Synthesizer, Unit 108, provides one or moreunmodulated “Carrier Wave” signal(s) to the QM. The quadrature-modulatedsignals are provided to the Transmit Amplifier (AMP). The amplifier maybe operated in a fully saturated mode, designated as Non-LinearAmplifier (NLA) or C-class amplifier or it may be operated in aLinearized (LIN or Lin) mode. Between the QM-109 and the Transmit AMP111 part of the “RF Head” an optional combiner 110 is shown. Combiners110 and/or post-AMP combiner 111 are optional units for Pilot Insert(PI) 1 and 2 designated as units 109 and 112, respectively. The RFhead's transmit AMP is connected to Switch (SW) and/or Combiner/Splitterdevice 113 and this 113 unit provides the signal for transmission to andfrom antenna 114. Instead of the aforementioned antenna a separate portcould be used for signal transmission or reception of “wired” systems.The Pilot Insert (PI) optional units may provide in band/or out of bandpilot tones for transmission. These tones could be used for fast androbust performance receiver demodulators and synchronizers. In thetransmit and receive sections Switch 113 or Switch or Combiner orDiplexer is connected to the receive Band Pass Filter (BPF) 115. Insteadof Switch 113 a Combiner/Splitter could be used. The received signalafter the BPF is connected to a Low Noise Amplifier (LNA) unit 116. Theoptional down-converter unit 117 receives its inputs from the FrequencySynthesizer and from the LNA and provides it to band-pass filter (BPF)118 for further processing. The entire down-conversion stage of thereceiver, including the receive section of the Frequency Synthesizer,mixers 117 and 118 are deleted for the so-called Direct Down-Conversiontype of receivers.

The quadrature demodulator (Quad Demod) may contain components such asAutomatic Gain Control (AGC), Frequency Tracking, Synchronization andPost Demodulation, Signal Conditioning including Symbol Timing Recovery(STR) circuits. The demodulated signal provided by demodulator unit 119is fed to the receive section of the IU 102. The receiver section of theIU 102 contains on lead 101 the output signal and output port.

In FIG. 1 b, an alternate, somewhat generic Transmitter (Tx)block-implementation diagram of this invention for Bit Rate Agile (BRA),Modulation Format Selectable (MFS) hardware, firmware and/or softwareimplementations, including optional single and multi-tone inserts at oneor more locations and optional ACM embodiments, is depicted. Severaloptional interface units prior to the processor unit 132 are illustratedin this figure. These units may perform the transmit interface functionsand the corresponding receive interface functions. Illustrative examplesfor the transmit interface functions are described. Unit 120 illustratesa Forward Error Correction (FEC) and/or a Differential Encoder (DE).Unit 121 is an interface for Frequency Division Multiplexing (FDM); forCollision Sense Multiple Access (CSMA) 122 would be used. For CodeDivision Multiplexes (CDMA) and its variations such as W-CDMA and B-CDMAand 3^(rd) generation CDMA 123 would be used. For time divisionmultiplexed (TDM) 124, while for Continuous single or multiple digitalor analog signals or Analog to Digital (A/D) converted signals unit 125is used. For Telemetry interface 126, while for Broadcast Signalinterface 127 is used. Unit 128 is suitable for Orthogonal FrequencyDivision Multiplexing (OFDM). For additional CDMA processing orinterface 129 could be used. Trellis Coded Modulation (TCM) basebandprocessing of the trellis encoder and corresponding optimal demodulationdecoding could be performed in 130. Unit 131 is reserved for “other”emerging applications. Unit 132 is the processor, including the BBP fora generic class of signal generators disclosed in this invention. Thebaseband “in-phase” (I) and quadrature phase (Q) baseband signalsgenerated by 132 are provided to QM 133. The QM unit receives an inputalso from the Carrier Wave (CW) generator also designated as LocalOscillator (LO) which could be part of the Frequency Synthesizer (FS)140. The quadrature modulated output signal in one of the optionalembodiments is combined with one or more Pilot Tones in combiners 134and/or 136. Signal Amplifier (AMP) 135 provides the amplified signal tothe Transmit and Receive Antenna 138 through switch or combiner/splitter137. For wired applications such as telephony, coaxial cable, fiber andother physical wired connections interface unit/amplifier 139 is used.Bit Rate Agile Clocks (BRAC) are generated and/or processed in 141. TheControl (CTL) signals are obtained from unit 142 while one or multiplerate or sub bit rate Sampling Signals (SAM) are generated and/orprocessed by 143.

In FIG. 2 an embodiment of a BRA integrated Base-Band Processor (BBP)with BRA post filters and BRA Cross-Correlated (CC) signals and ACMfilters for quadrature modulation is illustrated. New BRA, MFS, CS andRF frequency agile implementation architectures of this invention aredescribed in conjunction with FIG. 2. On signal leads 2.10 and 2.11 thein-phase input (Iin) and the quadrature phase input (Q_(in)) signals areprovided to switch units 2.14 and 2.15. On lead 2.12 a serial inputsignal (Sin) is illustrated. This signal is connected to aSerial-to-Parallel (S/P) converter and/or Splitter unit, combined orindividually also designated as “splitter” 2.13. Switching and/orcombining units 2.14 and 2.15 provide in-phase (I) and quadrature phase(Q) baseband signals to the BRA processor 2.16. This BRA processor alsoreceives a set of Clock, Control, and Sampling Signals C, CTL and SAM,commonly also referred to as set of C signals or merely “C” signal or“C” for clock. The I_(out) and Q_(out) signals are on leads 2.17 and2.18 and provide inputs to the QM. The QM 2.19 provides drive signal (S)to amplifier 2.20 which in turn provides amplified signals to switch orcombiner/splitter unit 2.21. Antenna 2.22 and/or interfaceport/amplifier 2.23 provide the signal to the transmission medium. Unit2.19 Frequency Synthesizer (FS) provides the Carrier Wave Signals to oneof the inputs of the QM and may provide one or more pilot tones to theQM and/or the output of 2.20 for combining the pilot tone(s) with thequadrature modulated signal.

In an alternative implementation of the baseband processor, in the lowerpart of FIG. 2, on leads 2.24 and 2.25 parallel input signals(I_(parin)) and (Q_(parin)) are provided, while on lead 2.26 a serialinput signal is provided. Units 2.27, 2.29, 2.30, 2.32, and 2.33 areswitching devices for the serial signal input and for the I and Qsignals. Unit 2.28 represents a signal splitter. Signals and units 2.10to 2.33 constitute some of signal processing components of this newarchitecture. These components have related structures to thedescription contained in Feher's prior art U.S. Pat. No. 5,491,457, Ref[P2]. However, significant structural and implementation differencesexist between the structures described in [P2] and the entire structureand embodiment of FIG. 2.

The architecture and embodiment of the Bit Rate Agile embodiment of theFIG. 2 processor includes BRA Baseband Processor (BBP) 2.34 withcascaded BRA Time Constrained Signal (TCS) response and Long Response(LR) signal generators. BRA post filters with Cross-Correlated (X)Cosine (C) and Sine (S) single or multiple processors and inverters(XCSI) are also part of the new structure. These structures, devices andarchitecture, including the new ACM filtering features available withthese new elements are part of this invention.

Description of an Exemplary Embodiment of a FQPSK Bit Rate AgileTransceiver

In this part of the detailed description of the invention the focus ison Quadrature Modulated (QM) four (4) state FQPSK systems. These fourstate systems have in general, in the sampling instants, in the I and Qbaseband channels 2 signaling states (for short “states”). In thebaseband I channel and in the baseband Q channel there are 2 signalstates. The architectures and embodiments of nine (9) state FQPSKsystems are essentially the same as that of 4 state FQPSK Transceivers.An exception is that the baseband I and Q Cross-Correlated and BRAsignal processors provide 3 level (state) baseband I and Q signals andresults in 3×3=9 state modulated FQPSK systems. Most implementations andembodiments and alternatives apply to multi-state (more than 4 state)Quadrature Modulated systems, described in later sections, such 9, 16 or64 or 256 state QM systems having 3, 4, 7, 8 and 16 states in theirrespective baseband channels.

FIG. 3 shows BRA baseband filters and processors preceded by a “basiccross-correlator (Xcor)” component of this FIG. 3. The basic Xcorcomponent, including the wavelet storage units and multiplexers may berelated to implementations described in the prior art Kato/Feher U.S.Pat. No. 4,567,602 [P5]. In some of the implementations of the presentBRA invention, for use in BRA post filtered/processed systemapplications, these basic Xcor elements are used with integrated and/orpost Xcor BRA filtered/processed ACM units. In some other embodimentsdifferent basic signaling elements including cascaded Time ConstrainedSignal (TCS) response and Long Response (LR) waveform generators and BRApost filters are used. In alternate embodiments of this invention thebasic Xcor is connected to Cross-Correlated (X) Cosine (C) and Sine (S)single Inverters (XCSI). In some other alternate implementations thebasic Xcor unit is not used. The combined embodiment and structure ofthe set of TCS and BCM signal generators, collectively designated asunit 3.11, combined with Multiplexers 3.13 and 3.14 and cascaded withthe BRA cascaded TCS response and LR filters and baseband post filters3.15 is different from that of the aforementioned prior art [P5].

FIG. 4 shows a trellis coded processor implementation architecture.Variations of this structure include one or more elements illustrated inFIG. 4. The shown structure is suitable for encoded signal generationincluding Differentially Encoded (DE) and/or Non-Redundant TrellisCoding (TC) of the Baseband Processed filtered I and Q signals.Alternately other Forward Error Correcting (FEC) devices are containedin unit 4.13. The architecture of this FIG. 4 is suitable for several2^(nd) generation FQPSK embodiments. Elements of this structure generateenhanced spectral efficiency LIN and NLA constant envelope andnon-constant envelope systems. BRA digital and analog implementations ofACM selectable parameter cross-correlators and several sets of transmitand selectable receive filters are included. Synchronous single datastream and asynchronous multiple data stream input processing has beenalso implemented with the shown structure of FIG. 4. Transmitters withand without preambles and trellis or other encoders and/or analog anddigital pilot insertion as well as multi-amplitude cross-correlatedsignals including 3 level 7 level and multi-level partial responsesignals are also generated with the “mix and match” flexible andinteroperable elements and structures.

On leads 4.11 and 4.12 the in-phase and quadrature phase input signalsIin and Qin are illustrated. Units 4.13 and 4.14 contain Digital SignalProcessors (DSP) or FPGA logic elements, or other readily availableprocessors. These processors perform functions such as Trellis Coding(TC) with or without redundancy, Differential Encoding (DE), DigitalSignal Mapping for TC or other logic and/or memory modifications of thebaseband signals. The Digital Processing Addition (DPA) unit 4.15 isprovided for additional optional DE, Digital Pilot Insertion, andaddition of Forward Error Correction (FEC) bits and/or symbols,including CRC and/or pi/4 rotation of the QM signal. Element 4.16 is aBit Rate Agile (BRA) Base-Band Processor (BBP) which includes post-BBPACM filters and/or post-Cross-Correlation (CC) filters and processors inthe I and Q channels or a shared filter for the I and Q channels. Theaforementioned functions and implementations could be used in thedescribed sequence, or in a permutation or combination or variation ofthe aforementioned sequence. The entire processor of FIG. 4 could beimplemented in one or more integrated steps without specific separationof the aforementioned functionality and/or implementation architectures.

One of the implementation alternatives of the Differential Encoded (DE)algorithm, used in some of the embodiments of this invention isdescribed in the following paragraphs. In FIG. 4 the optional DE is partof unit 4.13 and is used for FQPSK and in particular for a specificFQPSK embodiment, designated as FQPSK-B, Revision A1 as well as FGMSK,FMSK and FQAM Transceivers. A somewhat more detailed implementationblock diagram of entire Differential Encoder (DE)-Differential Decoder(DD) and corresponding Serial to Parallel (S/P) and Parallel to Serial(P/S) converters is shown in FIG. 18.

A Serial-to-Parallel (S/P) processor, such as processor 18.1 of FIG. 18,is inserted prior to the DE 4.13 of FIG. 4 or prior to the DE unit 18.2of FIG. 18. The Differential Encoder 4.13 Differentially Encodes (DE)the I and Q data streams. The baseband I and Q signals aredifferentially encoded as follows:I=D_(even) _(—) Q^(bar)Q=D_(odd) _(—) IIn an alternative implementation of the aforementioned DE the inversion“bar” is on I in the second equation instead of the Q of the firstequation (that is I-bar rather than Q-bar).I=D_(even) _(—) QQ=D_(odd) _(—) I^(bar)

In one of the FQPSK embodiments, designated as FQPSK-B, one of theimplementations has the I and Q data symbols offset by one bit time(Tb), corresponding to a ½ symbol time (Ts),i.e. Tb=Ts/2. D_(even) andD_(odd) are the even and odd input data bits. In alternate embodiments,the aforementioned offset is not used. In the case of Direct SequenceSpread Spectrum systems such as certain CDMA systems, instead of theoffset by 1 bit time the offset could correspond to 1 chip time or to apresetable time.

In FIG. 5 several Time Constrained Signal (TCS) response patterns areillustrated. TCS and/or TCS response patterns are illustrated, describedand defined with the aid of the synonymous terms “Signaling Elements(SE)”, “Signal Components” or “Wavelets.” The terms TCS, TCS processorand/or TCS response shall mean that the pulse response or alternativelythe impulse response of TCS processors is constrained to the length ofthe memory of TCS processors. This TCS response may have an impact onthe cascaded response of a TCS processor with that of subsequentfilters. TCS response cross-correlated and also TCS response signalpatterns without cross-correlation are illustrated in FIG. 5. These areused in some of the implementations of this invention as signalingelements or wavelets connected in cascade to Long Response (LR) filters.The term “Long Response filter” or “LR filter” means that the measurablepulse response and/or impulse response of an LR filter or processor islonger than the pulse response of the TCS response processor. In severalimplementations the LR filter is implemented by conventional filtersynthesis and design. Conventional filter designs include the design andimplementation of active and passive Bessel, Butterworth, Chebycheff,Gaussian, analog and digital filters and of hybrid analog/digitalfilters. In alternate LR filter designs Infinite Impulse Response (IIR)and also Finite Impulse Response (FIR) architectures are implemented.

The pulse response of TCS response processors is limited to the memoryof the TCS processor. In several embodiments the memory of the TCSprocessor and/or TCS cross-correlator has been between ½ and 3 symbolduration Ts intervals. The pulse response of LR filters is related tothe type of the selected filter, the roll-off and the 3-dB bandwidth (B)and bit rate duration (Tb). For example an 8^(th) order Chebychefffilter, having a BTb=0.5 could have a practical, measurable pulseresponse of more than 10 bit duration. A 4^(th) order Butterworth filterhaving a BTb=1 could have a practical pulse response of more than 4 bitduration, depending on the accuracy/inaccuracy of pulse response andresulting Intersymbol Interference (ISI) definitions and requirements.

The basic TCS signaling elements or “Wavelets” shown in FIG. 5 precedethe LR filters of the ACM embodiments. In alternate architectures thesequence of TCS and of the LR filters/processors is interchanged andalso used in parallel architectures for combined TCS and LR signalgeneration. The TCS signal elements or for short “Signals” or “wavelets”in FIG. 5 are described as follows: signal pattern 5.11 representspattern S1(t) of a Non-Return-to-Zero (NRZ) signal pattern. The NRZsignal pattern contains TCS signaling elements where the duration ofeach NWZ signaling element is constrained to one Time Symbol (Ts)duration. Signal pattern 5.12, also designated as S2(t), represents another one Time Symbol TCS response wavelet pattern. This TCS responsesignal could be generated by the “Superposed Quadrature ModulatedBaseband Signal Processor” (SQAM) Seo/Feher's U.S. Pat. No. 4,644,565,Ref [P4]. Another TCS pattern, designated as pureIntersymbol-Interference and Jitter Free (IJF) signal pattern, or pureIJF wavelet and pattern, based on Feher's U.S. Pat. No. 4,339,724 [P7]is signal 5.14 also designated as S4(t). It is a TCS pure IJF signalpattern corresponding to the alternate NRZ pattern example shown assignal 5.13 and designated as S3(t). Signals 5.15 and 5.16 designated asS5(t) and as S6(t) are in-phase (I) and quadrature phase (Q) half a TimeSymbol (Ts) time delayed NRZ baseband signals. In this designation halfa symbol duration corresponds to one Time Bit (Tb) duration that isTs=Tb/2. Signal patterns 5.17 and 5.18 corresponding to S7(t) and asS8(t) are additional illustrative examples of TCS signals. These TCSsignals represent in-phase (I) and Quadrature-phase (Q) basebandCross-Correlated (CC) or (Xcor) signal patterns which could be generatedby use of the Kato/Feher U.S. Pat. No. 4,567,602, Ref [P5]. These S7(t)and S8(t) cross-correlated TCS response signal patterns represent IJFencoded output signals having amplitudes modified from the peakamplitudes of IJF signals.

In FIG. 6 an implementation diagram with cascaded switched transmit (Tx)and receive (Rx) Low-Pass-Filters (LPF) in conjunction withcross-correlated and other non cross-correlated TCS response andcascaded LR processors is shown. These LR processors could beimplemented as separate I and Q LPF s or as an individual time-sharedLPF. The Transmit Baseband Signal Processor (BBP) including the I and QLPFs could be implemented by digital techniques and followed by D/Aconverters or by means of analog implementations or a mixture of digitaland analog components. External Clock and External Data Signals are usedto drive the S/P and the entire baseband processor (BBP). The BBP mayinclude a Differential Encoder (DE). The I and Q LPFs may be implementedas single filters (instead of cascaded filters). Modulation andDemodulation filters have been implemented and tested with intentionallyMis-Matched (MM) filter parameters. Some of the best performanceimplementations use Agile Cascaded Mis-Matched (ACM) architectures. LRfilters have been synthesized and implemented as phase equalized andalso as non-equalized phase response transmit and receive Bessel,Gaussian, Butterworth and Chebycheff filters. Bessel, Gaussian andButterworth and Chebycheff filters as well as other classical filtersare within the previously described and defined class of Long Response(LR) filters. These filters have a relatively long practical impulseand/or pulse response. The measurable practical pulse response of theaforementioned filters having an approximately BTs=0.5 design parameterextend to many bit durations. Here B refers to the 3 dB cut-offfrequency of the filter and Ts to the unit symbol duration. Fromclassical communications and Nyquist transmission theory it is wellknown that the theoretical optimal performance minimum signal bandwidthis defined for BTs=0.5. The LPFs in the I and Q channels, or the sharedsingle set of LPFs, implementations include Infinite Impulse Response(IIR) and Finite Impulse Response (FIR) filters.

In FIG. 6 on lead 6.3 a serial data stream is present. This signal isprovided to 6.4 a and the optional 6.4 b units for Serial-to-Parallel(S/P) conversion and a 1 bit duration (Tb) offset in one of theimplementations. In other implementations there is no offset delay 6.4 bin the embodiment. Some other alternate embodiments use a selectableoffset delay 6.4 b which is larger or equal to zero and smaller than theduration of approximately 200 bits. As stated the Offset logic is usedin certain embodiments, while in other architectures it is not present.The input signal or input signals are provided on leads 6.1 and 6.2instead of lead 6.3 in some of the alternative implementations of thisinvention. Unit 6.5 is a Base-Band Processor. Unit 6.5 may be clocked,controlled and sampled by signals such as C, CTL, and SAMP such asillustrated previously in FIG. 1 to FIG. 3. In this figure, FIG. 6, allof these clocking, control and sampling signals which could representmultiple rates and multiple leads are collectively or individuallyabbreviated simply as “C” and illustrated with an arrow near the letter“C.” Unit 6.5 in some of the embodiments performs the Time ConstrainedSignal (TCS) processing, waveform assembly and generation functions ofmultiple symbol TCS cross correlation and signal processing operations.The I and Q outputs of unit 6.5 are provided as inputs to the transmitset of LPFs designated as TX1 LPF-1 unit 6.6 and TXQ LPF-1 unit 6.11.This set of first LPFs could be cascaded with a second set of I and Qchannel LPFs units 6.7 and 6.13. Switch units 6.8 and 6.12 illustratethat the second set of LPFs could be bypassed and/or deleted in some ofthe embodiments.

The LR filter units, embodied by the first and second sets of I and Qare implemented as LPFs or alternately as of other types of filters suchas Band-Pass Filters (BPF) or High Pass Filters (HPF) or otherfilter/processor LR filter combinations. As stated previously, forseveral embodiments all of the aforementioned processors are BRA andACM, while for other implementations bit rate agility and/or ACM may notbe required. Units 6.9, 6.10, 6.14, 6.15 and 6.16 comprise a quadraturemodulator in which the I and Q modulators are 90-degree phase shiftedand in which a Local Oscillator (LO) is used as a Carrier Wave (CW)generator. Unit 6.17 is an amplifier that could be operated in a LIN orin a NLA mode. The output of amplifier 6.17 is provided on lead 6.18 tothe transmission medium.

In FIG. 6 at the receiving end, on lead 6.19, is the received modulatedsignal. Unit 6.21 is a BPF that is present in some embodiments while inothers it is not required. Alternatively the receive BPF could be“switched-in” or “switched-out” by switch 6.20. In some implementationsSurface Acoustic Wave (SAW) BBF were used to implement 6.21. Units 6.22,6.23, 6.24 and 6.25 embody a Quadrature Demodulator (QD) with acorresponding Local Oscillator (LO). The aforementioned LO representsfor some embodiments an entire Carrier Recovery (CR) subsystem while forother embodiments it is a free running LO. The set of LPFs 6.26 and 6.27are the embodiment of post-demodulation filters, while the second set ofLPFs 6.28 and 6.29 may be used to further enhance the spectralefficiency advantages or other performance advantages of designed ACMsystems. The second set of LPFs could be connected or disconnected byswitches 6.30 and 6.31 or entirely deleted. Unit 6.32 is the ClockRecovery (CR) and/or Symbol Timing Recovery (STR) system. For fast clockand/or STR, this unit is connected in some of the embodiments inparallel to the Carrier Recovery (CR) subsystem. In one of theembodiments of fast Clock Recovery (CR) systems, the parallelconfiguration embodied by units 6.37 and 6.38 is used for discretesignal clock generation. The discrete signal spike, in the frequencydomain, provides on lead 6.39 the clock recovery unit 6.32 with adiscrete spectral line signal which is exactly at the symbol rate or atthe bit rate. In this architecture unit 6.37 is a multiplier or anyother nonlinear device which has at its input the received modulatedsignal and the same received modulated signal multiplied by a delayedreplica of itself. The aforementioned delayed replica is generated byunit 6.38, a delay element. The receiver structure, shown in FIG. 6, isone of the many possible alternative receiver and demodulatorstructures. It is inter-operable compatible and suitable for BRA and MFSand CS reception, demodulation and/or decoding of the transmittedsignals embodied by means of the BRA and/or MFS and/or CS and/or ACMimplementation of the FIG. 6 transmitter embodiments.

Contrary to the teachings and wisdom of well established bandwidthefficient communication theory, of matched filter-optimal demodulationtheory and optimal data reception theories, in several embodiments ofthe current invention, substantially Mis-Matched (MM) modulator and thedemodulator filters have been implemented. Fundamental and pioneeringdiscoveries, regarding the cascaded pulse response of TCS response andof LR filter cross-correlated BRA implementations of modulator I and Qfilters and that of the implementations of “matched” and/orintentionally “Mis-Matched” (MM) demodulator filters are disclosed inthis part of the invention. In classical communication theory thedemodulation LPFs, and in fact the entire cascaded receiver anddemodulation filter responses are matched to the characteristics of themodulator and entire cascaded modulator and RF transmitter filters.Minimum bandwidth-maximal spectral efficiency, optimal performancerequires that the Nyquist minimum bandwidth theorems be satisfied forISI free and matched signal transmission reception. The intentionallyand substantially Mis-Matched (MM) transmit and receive filter designs,used in implementations of this invention lead to simplerimplementations than implied by communication matched filter theory andby Nyquist minimum bandwidth theories and to substantially improvedperformance for RF power efficient NLA transceivers. From communicationstheory, numerous books, referenced publications as well as from patentsit is well known that for “optimum” performance the cascaded filters ofthe modulator should be matched by the cascaded receive demodulatorfilters. For example, in a conventional bandlimited QPSK system, ifNyquist filters are implemented as “raised cosine filters”, then thebest “optimal” performance is attained if the cascaded transmit andreceive filters have a raised cosine transfer function and the filteringis equally split, i.e. “matched” between the transmitter and receiver.For pulse transmission, such as filtered NRZ data an aperture equalizer,having an wTs/sin(wTs) frequency response is used in theoretical optimaltransmitters, prior to the implementation of the transmit matchedfilter. Specifically, based on Nyquist transmission and filter theories,combined with matched filter receiver theories the 3 dB cut-offfrequency of an optimal minimum bandwidth transmit filter, used as abaseband I or Q channel filter, in a QPSK system equals ½ of the symbolrate or alternatively ¼ of the bit rate. The 3 dB bandwidths of themodulator and demodulator filters of the “theoretical optimal”bandlimited QPSK system are matched. The 3 dB bandwidth of thetheoretical optimal system it is the same for the modulator filter andfor the demodulator filter. If these filters are implemented bypre-modulation LPFs and post-demodulation LPFs then the aforementionedtheoretical bandwidth corresponds to BTs=0.5. This value corresponds toBTb=0.25, where B is the 3 dB bandwidth of the respective filters, Ts isthe unit symbol duration and Tb is the unit bit duration.

Contrary to the teachings of the aforementioned optimal performancematched filter modulation demodulation theory, we disclose theimplementation of demodulator architectures and embodiments with“Mis-Matched” (MM) filtering, and specifically for agile (bit rate)cascaded mismatched (ACM) implementations. The term Mis-Match (MM)refers to intentional and substantial MM between the cascaded 3 dBbandwidth of the I and Q demodulator filters and/or to the MM withrespect to the Nyquist theory stipulated bandwidth. with that of thecascaded response of the modulator I and Q filters. Alternateembodiments include MM pre-modulation baseband LPF and post-demodulationbaseband LPF designs as well as post modulation BPF transmitterimplementations and receiver pre-demodulation BPF implementations. Acombination of the aforementioned baseband and BPF designs has been alsoimplemented. The term “substantial” MM in a BRA architectures andembodiments such as shown in the alternate implementation diagrams inFIG. 6 or FIG. 7 or FIG. 10 to FIG. 15 and/or FIG. 25 or FIG. 30 impliestypically more than about 30% mis-match between the respective 3 dBcut-off frequencies of the transmit and receive filters, but this valueis exemplary and is not a limiting amount of mis-match.

One of the best known BRA implementations of FQPSK systems is designatedas the “FQPSK-B” family of transceivers. In this section, several bestembodiments of FQPSK-B Transceivers, operated in NLA systems aredescribed. The implementation of the FQPSK-B embodiment, described inthis section has BRA, CS and MFS architecture with substantially MMmodulation and demodulation filters. The modulator TCS responseprocessors cascaded with the LR filters and the demodulation filters areMis-matched (MM). In this FQPSK-B implementation a cross-correlationfactor of A=0.7 has been implemented between the I and Q baseband TCSresponse processors which are cascaded with the LR filters. The TCSwavelets and assembly of the TCS wavelets has been described in theKato/Feher patent Ref [P5]. A resulting I and Q cross-correlated datapattern of this implementation, at the TCS processor output and prior tothe BRA LR filters is shown in FIG. 5 as TCS data patterns S7(t) andS8(t) having a cross-correlation parameter A=0.7. The I and Q basebandsignal patterns are generated with several structures and elements,described as parts of this invention. The aforementioned TCS signalpatterns are available at the outputs of the following elements: in FIG.6 at the output leads of TCS 6.5, in FIG. 7 at the output leads of D/AUnits 7.8 and 7.11. In FIG. 8 at the output leads of units 8.7 and 8.8.In FIG. 10 b at the output leads of the D/A converter units 10 b.17 and10 b.18, and in FIG. 15 at the output leads of D/A converter (designatedas DAC's) units 15.7 and 15.14.

The aforementioned BRA, MFS, CS and intentional Mis-matched (MM)implementations and embodiments of four state FQPSK Transceivers,operated over NLA systems are applicable to Multi-State NLA systems,described in later sections.

In FIG. 7 one of the alternate implementations of Bit Rate Agile (BRA)transmitters is shown. The illustrated embodiment of the currentinvention uses a variation and alternative implementations of the “BasicCross-Correlator” (XCor) and post cross-correlation processors, asdisclosed in prior patents of Feher et al., with several originalembodiments described in conjunction with FIG. 7. In cascade with thebasic Xcor which implements TCS response processed cross-correlated orTCS not-cross-correlated signals is a second set of LR filteredprocessors. In cascade with the 2^(nd) processor and with Digital toAnalog (D/A) converters are pulse shaping bit rate agile LR filters,implemented as Low-Pass-Filters (LPF) or other type of filters in the Iand Q channels. As stated previously, BPFs, HPFs or other types ofprocessors/filters could replace the LPFs. On lead 7.1 is the inputsignal to unit 7.2, which implements S/P, DE and Gray encoding and/orother logic functions. Logic 7.3 is a cross correlator that is used tocross correlate I and Q signals. The duration of the cross correlationprocessor and implementation of the basic Xcor is selectable in thecurrent invention. It is selectable in a wide range, from zero, i.e. nocross-correlation to a fraction of a bit interval, and is adjustableand/or selectable up to many bits and/or symbols. In some of thealternate implementations of FIG. 7 the entire cross-correlation Unit7.3 designated as “Logic” is not used, that is, it is deleted from FIG.7, in the generation and assembly of the TCS response signals, providedby the signal generator set 7.4. In logic/cross-correlator 7.3 sixsymbol shift registers are shown for the I and Q channels. As stated inalternative embodiments the cross correlation is deleted. The basicsignaling elements, also referred to as wavelets, designated F1, F2, . .. , F16 are generated and/or stored by a set of signal generators orstorage devices designated as 7.4. The aforementioned storage units orwavelet generators are implemented in one of the alternate embodimentswith ROM and/or RAM chip sets and/or are part of a firmware and/orsoftware program. In certain embodiments a fairly large number ofwavelets are generated, i.e., a set of S0, S1, S2, . . . , S63 or evenmore wavelets are generated, while in other embodiments only 2 or 4signaling elements (wavelets) are used. In alternate implementationsinstead of generating and/or storing separate and distinct signals or“wavelets”, a very small number of wavelets is stored and theirinversions in terms of amplitude inversion and time inversions are used.In the embodiments of the current invention the elements are suitablefor BRA and ACM operation. Units 7.5 and 7.6 are designated as twomultiplexes and are embodied in some implementations as a singleintegrated multiplexer unit or more than one unit. In one of theembodiments all digital processors, including units 7.2 to 7.16 areimplemented as a single function and unit. Unit 7.7 is a secondprocessor and provides for optional additional cross correlation andamplitude limiting, also designated as signal clipping or Peak Limiter(PL). PL are implemented by clipping devices and/or other commerciallyavailable prior art nonlinear devices. Other conventional devicesdescribed previously in these specifications as well as in the prior artliterature embody units 7.8 to 7.13. Amplifier 7.15 provides the RFmodulated signal to the antenna 7.17 through a switch and/or combiner orsplitter 7.16.

In FIG. 8 a Quadrature Modulated (QM) Transceiver embodiment of thisinvention is shown. On lead 8.1 the input signal is provided to theoptional S/P and/or DE and/or Logic/Coding processor 8.2 a. Unit 8.2 bis an optional (Opt.) Xcor, designated in FIG. 8 as Xcor 1. Unit 8.2 bprovides I and Q signals through the optional offset delay, D1 unit 8.3or optional bypass switch 8.4 for further processing to units 8.5 and8.6. Units 8.5 and 8.6 implement one or more I and Q processingoperations of Intersymbol Interference (ISI) and Jitter Free (IJF)signals such as cross-correlated amplitude-adjusted IJF signals or otherTCS cross-correlated and/or non-cross-correlated signals includingbinary and multilevel SQAM signaling elements in cascade with LR filtersubsystems such as IIR and/or FIR processors which are operated in a BRAmode. Units 8.7 and 8.8 are D/A optional single shared D/A, or multipleD/A converters that provide signals to the second set of filters 8.9 and8.10. Analog, digital or hybrid hardware, software or firmware in unit8.11 implements an optional cross correlator. The I and Q output signalsof 8.11 are provided as baseband drive signals of the QM 8.12. LocalOscillator (LO) 8.13 provides the RF unmodulated CW to the QM. The QMprovides to Amplifier 8.14 a signal for amplification to antenna 8.15 orto the transmission medium. One of the embodiments has a verysimple/efficient implementation of the TCS response cross correlatedtransmit (Tx) processor with only 4 samples/symbol and only three (3)wavelets.

In FIG. 9 Peak Limited (PL) and other TCS response signal patterns ofthis invention as well as that generated by the prior art SuperposedQuadrature Modulated Baseband Signal Processor (“SQAM”) Seo/Feher's U.S.Pat. No. 4,644,565, Ref. [P4] are illustrated. Signal pattern designatedas 9.11 is a conventional prior art NRZ signal, while TCS pattern 9.12,if it is not cross-correlated with an other signal or not cascaded witha LR processor, represents a processed prior all SQAM generated basebandsignal wavelet pattern. If the TCS pattern 9.12 signal is connected toBRA one or more filters, including LR filters and/or ACM Processor's andit is part of I and Q cross-correlated signal generators and BRAprocessors, then it is a new implementation of this invention, having asubstantially enhanced bandwidth efficient signal. Signal pattern 9.12is reproduced as signal pattern 9.13. The dotted line of signal pattern.9.14 illustrates the TCS processor and cascaded Long Response (LR)filtered output sample signal pattern. The 9.14 LR filtered signalpattern is more spectrally efficient than the 9.12 TCS response signal,however it may contain more Intersymbol-Interference (ISI), more DataTransition Jitter (for short “Jitter”) and higher signal peaks than theTCS signal pattern. A PL or “Clipped” signal pattern 9.16 of thisinvention is illustrated with dotted lines adjacent to signal pattern9.15. Signal clipping and/or smooth gradual or soft PL or abrupt peaklimiting of TCS, and/or of TCS response processors cascaded with LRfilters and/or cross-correlated signals reduces the overall signalexcursion of the peak to peak amplitude variation of the I and Qsignals. It also reduces the envelope fluctuation of the I and Qmodulated signal. Applications include I and Q analog and digitalsignals, including Orthogonal Frequency Division Multiplexed (OFDM),clear mode non-spread spectrum as well as spread spectrum signals, suchas CDMA and also TDM or TDMA, CSMA and FDMA architectures.

A Cross-Correlated (Xcor) embodiment of this invention for a BRAarchitecture, and also suitable for ACM operation, is shown in FIG. 10a. It is suitable for several classes of FQPSK, FMOD, FGMSK, FMSK, FGFSKand FQAM implementations. The implementation includes Gaussian LPF andIntegrator and/or other filter processor as well as sin and cos look-upand/or other “look up tables”. BRA implementations and implementation ofTCS and LR filtered/processed Cross-Correlated (CC) baseband I and Qsignals and the corresponding Quadrature Modulator (QM) and RF amplifier(Amp) are included in the embodiment of FIG. 10 a. The aforementionedterm “look up tables” refers to TCS response wavelet storage and/orwavelet generation units. In alternate embodiments the sin and coswavelet generators and/or wavelet storage and readout units are replacedwith other wavelets than the aforementioned sin and cos functiongenerated wavelets. On lead 10 a.1 an input analog, digital or mixedsignal is provided to Unit 10 a.2 also designated as Filter 1. Thisfilter as well as other components of the transmitter areClocked/Controlled and/or Sampled by one or more 10 a.3 signals,designated for short by the letter C. Filter 1, 10 a.2 is a Gaussianshaped, Butterworth, Bessel or other filter or combination of filtersand processors and it is configured in LPF, BPF or HPF mode. Unit 10 a.4is a second signal processor. It embodies an Integrator or some TCSsignal shaping units and/or LR filter ACM implementation. Splitter 10a.5 splits the signal into I and Q signals and it may implement thesplitter and/or Serial to Parallel (S/P) converter and variations ofelements 2.10 to 2.15 and/or 2.24 to 2.33 of FIG. 2. The I and Q signalsprovided by the 10 a.5 splitter output are further spectrally shaped andlimited by the set of I and Q channel LPF's, designated as Filt 2I andFilt 2Q. These optional (Opt.) F2I and F2Q Bit Rate Agile Filter (BRA)elements 10 a.6 and 10 a.7 in FIG. 10 a are TCS and/or Long Response(LR) filters, where the term “Long Response” refers to the typicallylonger pulse and/or impulse response of the LR filters than that of theTCS filters. This cascaded TCS and conventional filter approach isapplicable for bit rate agile spectrum enhanced GMSK signal generations,also designated as FGMSK containing elements related to U.S. Pat. No.5,789,402 (for short '402); however, with the new single or multiple ACMCross-Correlated (X) Single or Multiple Cos (C) and Sin (S) and/orInverter (I) (for short “XCSI” and/or Peak Limiter (PL) processorcomponents of this new invention.

The second optional (Opt.) filter set is 10 a.6 and 10 a.7. This setprovides for processing of the I and Q signals provided by 10 a.5. TheXCSI unit 10 a.8 cross-correlates the aforementioned I and Q signalswith a C driven clock sample/control and provides for selectable amountof cross-correlation between the I and Q channels, including for across-correlation reduced to zero, i.e. no cross-correlation between theTCS and LR processed signals. In some of alternate embodiments of thisinvention there is no cross-correlation apparatus used. In units 10 a.9Iand 10 a.9Q optional additional filtering processing is implemented byoptional units F3I and F3Q. These units provide optional ACMfiltered/processed BRA cross-correlated or not-cross-correlated I and Qsignals to the Quadrature Modulator unit 10 a.10 which in turn providesthe signal of amplification to Amplifier (Amp) 10 a.11 and to outputport 10 a.12 for signal transmission or broadcasting. The output port isrepresented by connector 10 a.12.

In FIG. 11 a.1 a conventional Non-Return to Zero (NRZ) signal pattern isillustrated in 11 a.1. In 11 a.2 a modified Return to Zero (RZ) patterndesignated as Feher Return to Zero (FRZ) data pattern is shown. Both ofthe NRZ and FRZ signals are Time Constrained Signals (TCS). The FRZsignal 11 a.2 has an adjustable amount of Delay (D) for the forcedtransition instance from the logic state 1 to logic state 0. The FRYsignal is used in some of the TCS embodiments, 11 a.3 while in other TCSarchitectures FRZ is not in use. One of the advantages of FRZ signals isthat they may contain discrete spectral lines and have a more robustperformance in RF time dispersive or frequency selective faded systemsif used in conjunction with modulators/demodulators, including but notlimited to quadrature modems. The FRZ signal contains, in someembodiments, non-symmetrical or asymmetrical rise and fall times andeven pulse shapes. The TCS unit 11 a.3 has as its inputs one or moresignals as well as the “C” signals where C designates optional clocks,and/or sampling including over- and/or under-sampling signals and/orother control signals. The TCS processor in one of the embodimentsgenerates Non-Cross-Correlated (NCC) signals while in another embodimentCross-Correlated (CC) signals are generated in unit 11 a.3. In the caseof digital processing such as FPGA and/or ROM and/or RAM, digitalimplementations D/A converter(s) 11 a.4 are provided. In one of theembodiments a single D/A is implemented. The single D/A is providingtime shared or time multiplexed signals as I and Q inputs for QuadratureModulation (QM). Prior to QM an optional analog processor and/or filterwith possible BRA clocked operation, 11 a.5, is implemented. The QM 11a.6 provides BRA quadrature modulated signals to optional amplifier orcascaded amplifiers 11 a.7 which could operate in a linear mode orpartly linearized mode or in a fully saturated NLA power efficient mode.The aforementioned amplifier provides the signal to the Antenna 11 a.8or alternatively to the output port 11 a.9 for signal transmission.

An alternate embodiment is shown in FIG. 11 b. The implementation of aBit Rate Agile (BRA) pre-processor with single or multiple wavelets(also designated as) Signal Element (SE) storage and/or inverter and offiltered SE and ACM processors is illustrated. On leads 11 b.1 and 11b.2 respectively the Data Inputs (DI) and Clock (C) inputs are shown.Here the terms “Data Inputs” are synonymous with one or more digital oranalog or hybrid combined digital/analog signals having 2 or moresignaling states. The term “Clock” or for short “C” is synonymous withone or more clock signals, sampling and control signals as mentionedelsewhere in this disclosure. Pre-processor 11 b.3 stores I and Q SignalElements (SE) for BRA operation optionally controlled and/orclocked/sampled by the 11 b.2 and 11 b.4 signals/clocks. Thepre-processor unit 11 b.3 stores in one of the embodiments a largenumber of separate and distinct Signal Elements (SE), also designated as“wavelets”, and provides the selected SE in appropriate order to the 11b.5 single or multiple D/A converter(s). The D/A converter (S) providesto unit(s) 11 b.6 and/or 11 b.7 I and Q signals for further BRAoperation. Selectable filters designated as 1 to N and/or a bank offilters are used in one of the ACM embodiments of 11 b.6 and 11 b.7 andprovide the I and Q signals to ports 11 b.8 and 11 b.9.

In FIG. 12, including 12 a and FIG. 12 b predominantly analog componentsused in analog implementations and embodiments of this FQPSK and relatedtransceiver embodiments are shown. In addition to some of the activecomponents depicted in these figures, embodiments of the virtually samefunctions as those with analog components are implemented with passiveanalog components. The individual components are conventionaloff-the-shelf available components described in detail in prior artpublications including patents and detailed descriptions of thesecomponents are superfluous. In particular in FIG. 12 a analogimplementation components for cross-correlated and non-cross-correlatedBRA processed TCS and/or LR filtered and/or ACM signals are shown. InFIG. 12 a unit 12 a.1 is a “COS” that is cosine or sine wave generator.One of the well-known embodiments of 12 a.1 is a readily availableanalog Free Running Oscillator (FRO). Unit 12 a.2 is a “Direct Current”(DC) source while 12 a.3 and 12 a.4 represent an alternate COS sourceand a Square Wave Source, respectively. Even though COS and SIN sourcesare illustrated as separate components, embodiments include singlesource COS or SIN sources/generators which may be converted toTriangular Square Wave or other periodic or non-periodic sources. Theaforementioned signal sources or signal generators are connected in avariety of configurations such as the exemplary embodiments of FIG. 12 aand FIG. 12 b to components such as multipliers, amplifiers, switches,attenuators, inverting amplifiers and “DC shift” units 12 a.5 to 12 a.12for processing and providing signals for further processing includingselective switching or combining.

In FIG. 12 b another analog BaseBand (BB) embodiment using predominantlyanalog components is illustrated. Units 12 b.1 to 12 b.5 are signalsources such as previously described. Unit 12 b.6 is a previouslydescribed clock sample/controlled source abbreviated with the letter C.Unit 12 b.7 provides selection and/or combinations and permutations ofthe signal sources with amplitude adjustments of the aforementionedsignal sources which operate at the same rate in some embodiments andoperate in asynchronous non-related bit rates to each other in otherembodiments. Unit 12 b.7 provides one or more signals to standard andreadily available components such as 12 b.8 to 12 b.24. These componentsprovide inputs to the “select” or “combine” unit 12 b.25 which has atits input optional Control (CTL), Clock Bit Rate (CBR) and ControlSampling Time (CST) signals. One or more of these signals, by itself orin combination, may constitute multiple inputs and are synonymouslyfurther abbreviated and designated as “C” in several parts of thisinvention. Units 12 b.26 and 12 b.27 provide further signal processingfor the outputs of 12 b.25 and generate I and Q output signals.

In the embodiment of FIG. 13 a an implementation comprising mixed analogand digital circuit components is shown. On lead 13 a.1 the data inputor DIN provides two 13 a.2 signals for multiplexing control/logicprocessing and memory. In one of the embodiments, logic processing andmemory storage/processing converts the input signal to a Trellis Coded(TC) baseband signal. Control signals present on leads 13 a.3 serve asone of the inputs to the N by M (N×M) channel multiplexer and crosscorrelator and/or other TCS unit 13 a.6. Analog inputs of 13 a.6 couldbe generated by unit 13 a.4 and further processed by analog componentssuch as the set of components 13 a.5 of this particular embodiment.Units 13 a.7 and 13 a.8 provide additional digital timing bit rate andsample clock scaling, n/m rate division and miscellaneous logicfunctions embodied by standard logic gates and/or DSP and/or AnalogSignal Processing (ASP) components.

FIG. 14 shows an alternate design of the current invention including abank of switchable filters/processors with Linearized Phase or PhaseLinear (PL) and Not Linear Phase (NLP) Filters. Phase LinearizationComponents may be entirely deleted or, if included in thisimplementation, they may be switched in and out in this hybrid analogand digital transmit implementation of the transmitter. Unit 14.1 is theembodiment of a TCS processor for time constrained signals with orwithout cross correlation implemented for FQPSK or FGMSK or FQAM or FMSKor related BRA signals. The output(s) of the 14.1 TCS processor providesone or more signals to the optional D/A unit 14.2. The outputs of thesingle or multiple D/A or of 14.1, in case 14.2 is not used, areprovided to the cascaded I and Q filters and Switches 14.3 to 14.5.While the filters are indicated as LPF-1 to LPF-3, they have beenimplemented as low-pass, band-pass and high-pass filters providing theLR pulse response with or without phase equalization. These filters maybe designed for Agile Cascaded Mis-Matched (ACM) systems. An alternateembodiment comprises TCS unit 14.6 connected through optional D/Aunit(s) 14.7 to the bank of 14.8, 14.9 and 14.10 selectable or combinedswitched phase equalized or non-phase equalized parallel and/or cascadedunits LPF-1I, LPF-1Q to LPF-M1 and LPF-MQ filters. On output ports 14.11and 14.12 the processed I and Q signals of the aforementioned particularembodiment of this invention are available for further processing.

In FIG. 15 one of the predominantly digital BRA and ACM alternativeimplementations is shown. The embodiment of digital processor basedimplementation, followed by one or more D/A converters and BRA LongResponse (LR) filters are shown in the architecture of FIG. 15. Theblock diagram of FIG. 15 is an alternative embodiment of this inventionhaving a TCS signal generator in cascade with the 2^(nd) processor setof bandwidth spectral shaping LR filters. The LR filters are designatedas LPF. The embodiments include analog and digital LR filters includingcombination and selections between LPF, BPF and HPF and other analog ordigital filters. Data encoder 15.1 has at its input the “Input Signal”and the “C” signals. For simplification reasons of this and some otherfigures of this invention, the C signals are not always drawn on therespective figures and are not connected in the diagrams to all of thepossible inputs. Following data encoding, implemented by conventionaldigital and/or analog components and/or DSP and/or software or firmware,the signals are provided to an encoding/logic processor which couldcontain a Shift Register (SR) unit 15.2. A flag generator provides oneof the output bits to waveform select logic 15.4 for processing. Addressgenerators 15.6 and 15.9 provide addressing information to ROM units15.5 and 15.6. The ROM units are sampled and their content read out toDAC (Digital to Analog Converters) 15.7 and 15.14. The outputs of theDAC units are provided to units 15.8 and 15.15 for further filtering andproviding the output signals to ports 15.16 and 15.17, the ports for theI and Q signals, respectively.

In one of the alternate Field Programmable Gate Array (FPGA) basedimplementations of the TCS processor of FQPSK and FGMSK readilyavailable Xilinx Chip Model No. SC4005PC84-6 has been used. While in another embodiment Xilinx SC4000OFPGA was used. Other implementations usedAlterra and Intel devices. The wavelets used in one of the designs used16 samples for 1 symbol time Ts duration while in an other embodimentsused only 4 samples for 1 symbol time Ts. Each sample was encoded in oneof the implementations into 10 bits/sample with D/A converters having 10bit resolution, in other cases 4 bits/sample were used.

In one of the embodiments of FIG. 15, a data encoder, 4-bit shiftregister, I and Q waveform select logic, address generators based on ROMimplemented components, D/A converters, Clock and Sampling Generatorsand output latches are used. This particular design does not use ahalf-symbol physical delay component in one of the (I or Q) basebandchannels. Rather, the half-symbol offset between the I and Q channeloutput waveforms is obtained by appropriate waveform selectionprocedure.

In FIG. 16 “Wavelets” or Basic Signaling Elements (SE), for FQPSKdesignated also as “Wavelets” and/or “Signal Components” or merely“Signals” are shown. This FIG. 16 depicts TCS Wavelets for enhancedperformance FQPSK signal generation with a Cross-Correlation of A=0.7and in particular for TCS wavelets with only four (4) signaling elementsrequired for storage—suitable for high speed and integrated TCS andcascaded filter LR processing solutions. The appropriately assembled Iand Q cross-correlated TCS sequence is provided to BRA Long Response(LR) filtering of I and Q channels. Signals 16.1, 16.2, 16.3 and 16.4 inFIG. 16, also designated as S1, S2, S3 and S4, are illustrated acrossone symbol interval. For 4 signaling state systems one symbolcorresponds to two bits, thus Ts=2Tb in duration. The S1 to S4 signalscould be sampled and stored 4 times per symbol, i.e., 2 times per bit orat other sample intervals. The sampled signal wavelet values are storedin architectures containing memory devices. Signal element or “wavelet”S4, designated as 16.4, is simply a DC component in this particularembodiment. In some of the TCS embodiments S1 to S4 are cross-correlatedbetween the I and Q channels and have continuous derivatives at thesignal transitions, while in other embodiments the TCS signals are notCross-Correlated. Alternate embodiments have a more or lesser number ofwavelets than illustrated in FIG. 16.

An implementation of FQPSK baseband signal processor's taking advantageof inverse and symmetrical properties of its waveforms is described inthis section related to previously described figures and embodiments andin particular in relation to FIG. 15, FIG. 16 and FIG. 17. Instead ofthe storing all the basic entire or whole “Wavelets” of the basebandsignals, one of the implemented designs with ROM lookup table uses only3 “basic” wavelets. To simplify the implementation of cross-correlatedFQPSK baseband signals, we use the symmetry properties of the waveletsand the hold function for DC value. In the design there is a half-symboldelay between the I and Q channels so that the ROM contents in I and Qchannels are the same. FQPSK and FGMSK eye diagrams at the TCS outputand also at the cascaded TCS and LR filters are shown in FIG. 19. As aspecific design and teaching example the TCS generated eye diagram of anFQPSK signal with an A=0.707 cross-correlation parameter is shown inFIG. 19( c). Such an eye diagram is measured and/or computer generatedat the output of the embodiments shown in FIG. 10 or FIG. 11 and/orother disclosed embodiments, including at the outputs of the LR filtersin BRA operation of the implementation of FIG. 11, provided that the D/Aconverter has a good resolution accuracy e.g. 8 bits/sample and that theLR filter units 15.8 and 15.15 of FIG. 15 have a considerably highercut-off frequency than the inverse of bit rate. It can be seen that inthe FQPSK eye diagram of FIG. 19( c) there are 10 kinds of waveforms orwavelets during half-symbol duration (from 0 to 8 or 0 to Ts/2). Thus itis possible to pre-calculate the waveforms directly from the input data.FIG. 15 shows one of the implementation architectures of the designedFQPSK baseband processor using this approach. In FIG. 16 the 3half-basic waveforms needed to generate all the possible TCS responsewavelets for FQPSK are shown. Based on the symmetry and inverseproperties of the whole set of the wavelets illustrated in FIG. 19( c),instead of 10 only 3 waveforms plus a DC value (holding function for DCvalue) are required. In this design, 4 samples/symbol (or 2 samples perbit) are used to implement the waveforms of FQPSK.

In FIG. 17, BRA “Wavelets” for FGMSK are shown. These Gaussian waveletsare suitable for smaller size memory implementations, for bit rate agileGMSK signaling having a BTb=0.5 parameter. This FIG. 17 depicts TCSWavelets for use in LR filtered enhanced performance reduced spectrumBRA systems. Only four (4) signaling elements are required in thisembodiment of the TCS part of the FGMSK processor. With the embodimentof FIG. 15 and other alternate digital and/or analog embodiments thesesignals can be easily generated even at very high bit rates. Theappropriately assembled I and Q cross-correlated TCS sequence isprovided to BRA Long Response (LR) filtering of I and Q channels. InFIG. 17 the signals 17.1, 17.2, 17.3 and 17.4, also designated as S1,S2, S3 and S4, are illustrated across one symbol interval. For 4signaling state systems one symbol corresponds to two bits. Thus, Ts=2Tbin duration. The S1 to S4 signals could be sampled and stored 4 timesper symbol, i.e., 2 times per bit or at other sample rates. The sampledsignal wavelet values are stored in architectures containing memorydevices such as the previously disclosed ROM based embodiments. TheFGMSK signal shapes have different shapes from the S1 . . . S4 signalsillustrated in FIG. 16. In some of the embodiments for use in FGMSK theS1 to S4 wavelets are cross-correlated between the I and Q channels. Inother embodiments for FMSK the signals are not cross-correlated, or havedifferent cross-correlation algorithms and embodiments than in FGMSKand/or have continuous derivatives at the signal transitions. Alternateembodiments have a larger or smaller number of wavelets than illustratedin FIG. 16 and in FIG. 17.

In FIG. 18 Differential Encoding (DE) and Differential Decoding (DD) ofFQPSK and FGMSK is shown. A difference between the DE of this bit rateagile FGMSK encoder from that of conventional GMSK is in the algorithmdifference of these two DE and Corresponding Differential Decoding (DD)embodiments. The new DE for FGMSK is fully compatible and interoperablewith conventional OQPSK; the DE of prior art bit rate agile GMSK is not.

Eye diagrams shown in FIG. 19 are hardware measured and computergenerated diagrams for Cross-Correlated BRA signals at variousmeasurement/display points. The eye diagrams of DE prototype BRAapparatus transmit signals are presented. In FIG. 19( a) FGMSK eyediagrams of I and Q baseband signals with BTb=0.3 are shown prior to theLR cascaded performance enhancement I and Q filters. In FIG. 19( b) eyediagrams of I and Q baseband signals of an FQPSK transmitter, operatedin a BRA mode having a cross-correlation parameter A=0.7 after the BRAprocessor LR filters (I and Q filters) are shown. In this case the LRfilters have a relatively high cut-off frequency relative to the symbolrate. In FIG. 19( c) an FQPSK computer generated eye diagram isillustrated. The I channel eye is shown, for a Xcor. Parameter A=0.7prior to LR filters. The eye diagram contains only four (4) basicwavelets and represents a TCS eye pattern In FIG. 19( d) hardwaremeasured FQPSK eye diagrams of I and Q signals are shown for a BRAoperation displayed after the LR filters of the I and Q Channels. Thisparticular FQPSK is designated as an FQPSK-B and it has a Xcor parameterA=0.7 followed by I and Q post TCS Low-Pass Filters having LRcharacteristics. In FIG. 19( e) the measured vector constellation of anFQPSK-B signal after the LR filters processors of an implementedprototype system is shown.

Description of Multi-State FQPSK, FQAM, FGMSK and FMSK

In this section of the detailed description of this invention the focusis on Quadrature Modulated (QM) multiple signaling state (for short“state”) systems with more than 4 signaling states of the QM signal andmore than 2 states in the respective I and Q baseband channels. In theprevious section the focus was on the description of QM four (4) stateFQPSK systems. These four-state systems have, in general, in the I and Qbaseband channels 2 signaling states (for short “states”) in the Ichannel and 2 states in the Q channel. Most implementations andembodiments of the 4-state systems apply to multi-state (more than4-state) Quadrature Modulated systems, described in this Section, suchas 9, 16, 49 or 64 or 256 state QM systems having 3, 4, 7, 8 and 16states in their respective baseband channels. The technologies andembodiments described for the multiple state systems are also applicablefor the implementations and embodiments of 4-state systems. Forward andBackward COMPATIBILITY and/or interoperability between the 4-state and,more than four-state, multiple state FQPSK and Feher's QuadratureAmplitude Modulation (FQAM) and multi-state FGMSK and FQPSK systems is adefinitive advantage in new product developments. Additionally some ofthese systems are also backward compatible with the previously patentedFeher BPSK (for short FBPSK and FMOD) systems, such as disclosed in [P1]and [P2] and the references in the aforementioned U.S. Patents and citedreferences.

A multi-state QM architecture for 4 or more than 4 states, designated asFQAM is illustrated in the embodiment of FIG. 20. In this implementationblock diagram of an FQAM multi-state Cross-Correlated BRA Transmitter asingle RF Amplifier operated in fully saturated NLA mode or Linearized(Lin) mode of operation is used. The Input Signal is connected to anoptional Encoder unit 20.1. This unit, if used includes logic processingand Encoding functions such as Trellis Coding or CRC or FEC or DE orGray Coding, Serial to Parallel conversion and/or other digitalprocessing functions. In one of the embodiments an optional S/P (Serialto Parallel) converter, unit 20.2 is included to process the signalsreceived from 20.1. The Signal Mapper 20.3 maps the binary signals from2 states to M states (levels) and includes in some of the embodimentsCross-Correlation between the binary and/or between the convertedmulti-state signals. The outputs of the Signal Mapper TCS unit 20.3 areconnected to D/A converters 20.4 and 20.5. The D/A Cross-Correlated BRAsignals are TCS multilevel Cross-Correlated signals with variable and/orpresetteable Xcor. In alternate embodiments no cross-correlation betweenthe I and Q signals is implemented. The D/A outputs are fed to 20.6 and20.7 filters, indicated in the drawing as LPF. These filters are LRfilters and are implemented with IIR digital or IIR analog filters or acombination of conventional analog or digital filters. The BRA signals,which have been Cross-Correlated and are TCS in cascade with LR signalsare provide to the inputs of Quadrature Modulator (QM) 20.8. The QM hasan unmodulated Carrier Wave input from unit 20.9. The QuadratureModulated signal is processed by an optional “Roofing Filter” to removehigher order spurious components and is fed to amplifier 20.12 and toantenna 20.13 or output port 20.14. An optional Pilot Tone or MultiplePilots are added to the RF signal by Pilot Generator/adder 20.11.Combining adding of pilot signals is achieved by hardware combiners orby adding Unmodulated signals over the air through a separate antenna.

A 16 state FQAM embodiment is illustrated in the implementationarchitecture and block diagram of FIG. 21. This architecture can beextended and/or modified to 64 state or to other larger or smallernumber of signaling states. In simple terms NLA Quadrature Modulatedsystems are generated by FQPSK or FGMSK or FMSK type of 4 state QMembodiments (for short the generic term “FQPSK” is also used) asdescribed in earlier sections of this invention. If two NLA four stateRF signals are combined than a 16 state NLA signal is obtained. If threeNLA four state RF signals are combined than 64 state NLA signal isobtained. If four NLA four state RF signals are combined than a 256state NLA signal is obtained and the number of signal states can befurther increased by the aforementioned extension of the multiple signalcombining process. The term “combining” or “Combiner” or RF signaladdition in FIG. 21 is accomplished by an RF hardware combiner. Off theshelf, readily available components, such as Hybrid Microwave Combiners,are suitable for hardware RF combining. On lead 21.1 in FIG. 21 an inputNRZ data signal is provided to the input ports of the Serial/Parallel(S/P) Converter 21.2. Four parallel data signals, designated as 11, Q1and 12, Q2 are provided to FQPSK (or FGMSK or FMSK) modulators 21.3 and21.4. One of the embodiments and/or implementations of FQPSK previouslydisclosed in the detailed description of this invention implements 21.3and 21.4. The FQPSK signals are provided to Optional (Opt)preamplifiers, which operate in Linearized (LIN) or NLA mode. The HighPower Amplifiers (HPA) 21.7 and 21.8 provide the RF amplified modulatedsignals to RF Combiner 21.9, which in turn provides the NLA combined16-state signal to the output port 21.10.

In Seo/Feher [9] and [P4] and [6; 8] prior art references,implementation architectures of NLA systems operated in 16-QAM, 64-QAM,9-QPRS, 81-QPRS and other quadrature modulated systems have beendescribed and/or referenced. The aforementioned prior art does notinclude NLA cross-correlated, filtered and bit rate agile NLA systemsfor QAM disclosed in this invention, and in particular related to thediscussion of FIG. 21 and FIG. 22. Based on these Feher et al.references, one of the embodiments, is related to the architectures ofFIG. 21. To obtain a 16-state QAM with cross-correlated FQPSK signalsthat are NLA through HPA1 and HPA2, the RF amplifiers and RF combinerare adjusted to have an RF combined output fed to RF combiner, unit21.9, which provides the RF combined output 21.10. The RF combinedoutput power generated by unit 21.7—HPA1 is 6 dB higher than the RFpower provided to output 21.10 by HPA2 designated as unit 21.8.

FIG. 22 is the implementation architecture of “Over the Air Combined”FQAM signal generation by implementing 2 or more FQPSK and/or FGMSK typeof signals. In FIG. 22 instead of the use of a hardware embodied RFCombiner to combine the HPA1 and HPA2 signals, the output signals ofHPA1 and HPA2 are fed to two separate antennas and transmitted aswireless signals “over the air.” In this architecture RF Combining isachieved “Over the Air” that is the RF signals are transmitted over awireless medium and combined in the receiver antenna.

The baseband processor, quadrature modulator and signal amplifiersarchitecture of the NLA signals, “Over-the-Air Combined,” is closelyrelated to that of FIG. 21 which uses the hardware RF combiner. Theinput NRZ data on lead 22.1 is provided to a Serial/Parallel (S/P) unit22.2 which provides 4 parallel signals two FQPSK quadrature modulators,units 22.3 and 22.4. The quadrature modulated signals are provided toamplifiers, and in particular, to optional NLA 22.5 and to High PowerAmplifier (HPA) 22.7 in the upper part of the figure. HPA-1, unit 22.7provides the RF modulated signal to antenna 22.9 or to an output port22.11. In the lower branch the FQPSK modulator, unit 22.4, provides thequadrature modulated signal to optional NLA 22.6 and to HPA-2, unit22.8. The amplified signal of 22.8 is provided to antenna 22.10 and/orto output port 22.12.

To obtain a 16-state QAM with cross-correlated FQPSK signals that areNLA through HPA-1 and HPA-2, the RF amplifiers and RF combiner areadjusted to have an RF combined output fed to antenna 22.9 and/or outputport 22.11, 6 dB higher in power than the RF power provided to antenna22.10 and/or output port 22.12.

FIG. 23 shows the embodiment of an Orthogonal Frequency DivisionMultiplex (OFDM) type of embodiment with FDM signal combining of anumber of FQPSK type of Lin or NLA signals. In one of the embodiments ofthis invention RF Combining is implemented by hardware RF componentswhile in an alternative implementation the RF combining is implementedwith “Over the Air Combined” signals. If multiple antennas are used,then this architecture is also known as an “Antenna Array” (AA)architecture. Lead 23.1 containing the Input Data (ID) is provided to aSerial-to-Parallel (S/P) converter, unit 23.2, having M set of outputswhere one output set constitutes a separate I and Q signal or a serialdata stream. The S/P converter 23.2 may contain an optional crosscorrelator (Xcor). As the input signal is S/P converted, the data rateson the 1, 2, . . . , M input leads to the bank of FQPSK modulators 23.3are at an M times reduced data rate compared to the input data rate. Asan illustrative example of this architecture, if the input data rate isFb=10 Mb/s and there are 100 FQPSK modulators (M=100), then the bit rateof individual FQPSK modulators is 10 Mb/s: 100=100 kb/s. The aggregatetransmission rate of such a system is not changed. The aforementionedparallel data are provided to M modulators designated as FQPSK.1,FQPSK.2, . . . , FQPSK.M, collectively referred to as unit 23.3. Thesemodulated signals are provided to a set of RF amplifiers, 23.4 andoptional RF Switches units designated as 23.6. The M amplified signalsare provided in one of the embodiments through RF combiner 23.7 to asingle antenna “Ant.C” unit 23.8 or port 23.9. In an alternate use the Mmodulated and amplified signals at the outputs of RF amplifiers 23.4 areprovided to the antenna array 23.5 designated as Ant.1, Ant.2 . . . toAnt.M. If the Antenna Array architecture is used, then the M signals are“Over the Air Combined” signals. If a hardware RF combiner 23.7 is usedwith a single antenna 23.8 or output port, then the architecturerepresents a hardware combined embodiment. An advantage of the “Over theAir Combined” architecture is that all RF amplifiers within the set 23.4may operate in fully saturated NLA power efficient mode.

FIG. 24 is an alternate Antenna Array and RF Combining implementationarchitecture of multiple FQPSK type of signals. This figure illustratesmultiple TCS and/or filtered BB processors connected to an antenna arrayand/or RF combiner. Input lead 24.1 contains an input signal or amultitude of signals, which could comprise analog signals, digitalbinary or digital multilevel signals, or other signals, for short “InputSignal.” Unit 24.2 receives the input signal and processes it with a TCScontaining optional LR filters and/or a 1^(st) set of cross correlators(Xcor) for FQPSK signals. A Bit Rate Agile clocked bank of filters 24.3receives the signals of 24.2, processes them and provides to unit 24.4which is an optional 2^(nd) set of Xcor and/or TCS with LR responsefilters and/or Peak Limiter (PL) devices. The signals from 24.4 areprovided to Quadrature Modulators 24.5 and to a bank of amplifiers 24.6in the figure illustrated as FA1 to FA3. The RF amplified signals areprovided to antenna array 24.8 or to RF combiner 24.9 through the set ofRF switches 24.7. The RF combined output is provided to antenna 24.10 oralternately output port 24.11.

In FIG. 25 an alternate implementation of encoding and signal mapping ofFQPSK, FQAM and FGMSK signals is shown. In the embodiment of FIG. 25 atrellis coded implementation with an appropriate signal mapping fortrellis coded generic FQPSK without the need for redundancy is shown. Inaddition to trellis coding other coding algorithms are suitable for theshown implementation architecture. The aforementioned other encodersinclude non-redundant Differential Encoding (DE), Gray encoding,encoding conversion of NRZ signals into RZ or FRZ signals or Manchesteror other sets of signals. Encoding for error correction and detectionmay require the insertion of redundant bits and Forward Error Correction(FEC) encoders such as Block Encoders including Reed-Solomon, BCH,convolutional, CRC and other encoders are among the illustrativeembodiments of encoders suitable for FIG. 25. The specific trellisencoder schematic diagram and signal mapping part of FIG. 25 is based onSimon/Yan's published reference [22], which includes logic/encoding 25.3and signal mapping 25.4. The aforementioned reference does not discloseembodiments for TCS and LR filtered changeable amount of crosscorrelated and of filtered Bit Rate Agile (BRA) and/or Peak Limited (PL)implementations and of Agile Cascaded Mis-Matched (ACM) filtered systemshaving single or multiple I and Q readout tables with compressed memoryelements.

In FIG. 25 trellis coded TCS and LR filtered FQPSK signals having BRAapplications with or without Cross-Correlation and with and without PeakLimiting (PL) circuits in the I and Q channels are shown. The digitalprocessing parts of the trellis or other encoder precede the additionalTCS and LR and other processors in the implementation embodiment of FIG.25. In alternate embodiments the digital encoder with and withouttrellis coding is integrated in one block and one function with parts orall of the TCS and LR blocks or the order of processing is changed.

Illustrative Performance of Exemplary System

The performance of illustrative and some of the “best-illustrative”embodied FQPSK and related BRA systems of this invention is highlighted.This invention includes numerous embodiments and has a large class ofsubsystems and implementation details as well as designations. For thisreason the term “best-illustrative” is used herein. For some users anddesigners “best” means the narrowest possible spectrum at a PowerSpectral Density (PSD) of about −20 dB to meet certain FCC mandatedrequirements, while for other users the “best” PSD is defined at −70 dBfor others “best” refers to best Bit Error Rate (BER) performance in anAdditive Wide Gaussian Noise (AWGN) operated environment, while for someothers the “best” or “optimum” BER=f(Eb/No) performance is most desired.Other categories of the term “best” could mean fastest synchronizationor re-synchronization of a receiver/demodulator or “best” BERperformance or smallest number of outages in an RF delayspread-frequency selective faded environment with the “best,” that is,fastest and highest performance adaptively equalized demodulators. Alarge category of designers and users of this invention might define“best” as the lowest cost commercially available equipment from numeroussources having the highest spectral efficiency simultaneously with thesmallest size for a certain RF power and the “best,” i.e., maximal BitRate Agile (BRA) flexibility and/or interoperability and compatibilitywith previous generations and implementations of FQPSK and/or of other“legacy” systems. For the aforementioned reasons the term “bestillustrative” is used in the performance attained with some of theaforementioned “best” implementations.

As an illustration of one of the “best illustrative” NLA spectra, thePower Spectral Density (PSD) of NLA wireless/telemetry system isillustrated in FIG. 26. The multiple data links in these telemetrysystems are operated at specific bit rates of 13 Mb/s rate per link. Thespectra shown in FIG. 26 are typical for spectral usage in the U.S. forthe U.S. Government-authorized band of 2200 MHz to 2290 MHz forGovernment applications. In FIG. 26 one or more of the modulated andreceived signals is shown to be in the 18 dB to 20 dB lower than that ofadjacent signals. In “real-life” systems it is often the case that thedesired signal power is about 20 dB lower in power than that of theadjacent signals. With telemetry standardized filtered PCM/FM systems,at the aforementioned 13 Mb/s rate, 3 data links can be usedsimultaneously. During the 1990s, filtered PCM/FM systems have beenextensively used. With emerging FQPSK systems and in particular DraftStandardized FQPSK-B systems the number 13 Mb/s rate links is doubled to6, while with 16-state FQAM, also designated as FQPSK.2.4 the number of13 Mb/s links is quadrupled (over that of standardized PCM/FM) to 12data links. Thus the spectral efficiency of FQPSK is double that ofstandardized filtered PCM/FM and the spectral efficiency of FQAM with 16states and operated also in an NLA mode is quadruple that of thestandardized filtered PCM/FM systems.

From FIG. 26 illustrated spectra it is noted that the Integrated Powerfrom the adjacent channels falling into the desired channel may have asignificant impact on the performance of the desired channel. In thereferenced publications and patents the importance of IntegratedAdjacent Channel Interference (ACI) is highlighted and described. Forthis reason the Power Spectral Density (PSD) and the ACI of Linearly(Lin) and of NLA transmitted FQPSK and GMSK signals is shown in FIG. 27.The PSD and Integrated ACI of hardware measured prototype FQPSK-B in aNLA transmitter and of a BRA linearized (Lin) transmit FQPSK isillustrated in the upper part of FIG. 27. In the lower part of FIG. 27the Integrated Adjacent Channel Interference of FQPSK systems with thatof GMSK systems is compared. In this figure “W” denotes the frequencyspacing between Adjacent Channels. In case of GMSK which has beenimplemented with BRA embodiment described in this invention anddesignated as FGMSK having a Gaussian filter and bit duration (Tb)product of 0.25 has been used in the computed ACI result shown in FIG.27. The computed ACI of a BTb=0.25 filtered system with considerablysteeper filters than that of conventional Gaussian receive filters wassimulated by JPL/NASA. The ACI curves for FQPSK-B and FQPSK-D1 as wellas FQPSK-Lin are also included. Note from the results the significant(approximately 2 to 1 at −60 dB) spectral ACI advantages of FQPSK overthat of GMSK systems.

Multi-state FQAM systems such as 16-state FQAM designated also asFQAM-16 and FQPSK.2.4 (the first number in this latter abbreviationindicates 2^(nd) generation FQPSK while the 2^(nd) having 4 signalingstates per I and per Q channel). In NLA systems have approximatelydouble the spectral efficiency of the already-spectrally-efficient FQPSKsystems including BRA systems operated in an FQPSK-B mode disclosed inthis invention. The abbreviation “FQPSK-B” is a designation forcross-correlated FQPSK systems having an embodiment in which the TCScross correlators are cascaded with LR filters operated in a BRA mode.The substantial spectral saving attained by 16-state FQPSK systems overalternative conventional 16-QAM pre-RF amplification filtered systems isillustrated by the computer-generated results shown in FIG. 28.

In FIG. 28 the ACI results of illustrative FQPSK.2.4 (also designated asFQAM-16) state systems, obtained after fully saturated NLA, are comparedwith that of NLA conventional pre-modulation filtered 16-state QAMsystems. The spectral efficiency advantage of the illustrated FQAM ismore than 200% over that of prior art QAM at −30 dB and FQAM has an evenmore significant spectrum advantage in terms of ACI at the critical −40dB to −60 dB range. High performance systems require a robust BERperformance in Additive White Gaussian Noise (AWGN) and also otherinterfering and noise environments. A frequently-used performanceindication is the BER performance as a function of the available Energyof a Bit (Eb) to Noise Density (No) ratio. For RF power efficient aswell as spectrally efficient systems having a robust BER, the NLA systemperformance in terms of BER=f(Eb/No) is specified for numerous systemapplications. In particular, in FIG. 29 the performance of FQPSK ishighlighted.

FIG. 29 The BER performance, in terms of BER=f(Eb No), of prototypemeasurements and Computer design/software generated data of FQPSKsystems illustrated that NLA FQPSK is within about 0.5 dB of the idealtheoretical LIN amplified QPSK systems. Simple bit by bit detection aswell as trellis decoding without redundancy has been used. In FIG. 29curve (a) represents an ideal theoretical linearly (LIN) amplifiedsystem, while curve (b) represents a BER optimized NLA-FQPSK, and curve(c) represents a BRA and MM hardware-prototype measured FQPSK-B systemperformance, prior to optimization.

Detailed Description of Exemplary Embodiments of Receivers and ofDemodulators

Receivers and Demodulators are described in this section. Signalreception, adaptive equalization, demodulation, fast and robustsynchronization, bit recovery, Non Redundant Error Detection (NED),online in service monitoring, Non Redundant Error Control (NEC) and newarchitectures and embodiments for Bit Rate Agile (BRA) and NLAcross-correlated FQPSK, FQAM and related systems are disclosed in thispart of the invention.

In FIG. 30 a generic receiver and demodulator is shown. In this FIG. 30demodulation of FQPSK type of signals, by using quadrature demodulationstructures such as QPSK, QAM and OQPSK demodulation, enhanced by thearchitectures of the embodiment, illustrated in this figure isaccomplished. Fast signal acquisition is attained and numerous otherperformance enhancements are achieved by the disclosed genericembodiment of this demodulator. Input port and lead 30.1 obtains amodulated signal from a receive input port. The optional BPF unit 30.2is provided in some of the embodiments in which excessive strengthout-of-band/or adjacent channel signals degrade the performance of thesubsequent circuits if this “protection” or “roofing” BPF is notpresent. Automatic Gain Control (AGC) circuit 30.3 operates in a linearor in a non-linear (NL) mode. For some applications an advantage ofFQPSK and FQAM type of signals is that they are suitable for nonlinearAGC operation. From the prior art it is well known that nonlinearlyoperated AGC circuits such as “hard-limiters” and “soft-limiters” havefaster AGC operation than their linear counterparts and in certainenvironments nonlinear AGCs outperform linear AGCs and in particularregarding “weak signal suppression” and also lead to reduction ofCo-Channel Interference and of Inter-Symbol Interference (ISI) caused bycertain RF delay spread—frequency selective faded environments. Theoptional AGC provides the quadrature demodulator 30.7 and follow-upsignal processors including LPFs 30, Analog/Digital (A D) 30.10, SymbolTiming Recovery (STR) 30.6, Adaptive Equalizer (AE) which could be ablind equalizer 30.11 and digital logic circuits decoders 30.12 to 30.17the signals for demodulation and decoding. Additional signal processorswhich enhance the performance of the receiver/demodulator of thisinvention include one of the following: switchable changeable Carrierand Symbol Recovery unit 30.18, Pilot Extraction for unambiguous carrierrecovery, fast AGC, and Automatic Frequency Control 30.19. Non-RedundantError Detection (NRED) unit 30.20, Diversity Switching CombiningController 30.21 and Pseudo-Error On-Line Monitor 30.22 units are alsoincluded.

FIG. 31 is a demodulator architecture also known as a “BlockDemodulator” and/or digital demodulator or software demodulator. Thisimplementation/embodiment is used for software or firmware or hardware,or combined hybrid implementations of this invention. On receiveport/lead 31.1 the received RF signal or the received RF signaldown-converted to a convenient IF frequency or to the baseband frequencyrange is shown. Unit 31.2 is an optional “roofing” BPF or other filterto protect the front end of the A/D 30.3 unit from out-of-band signaloverload. The A/D provides serial or parallel signals for furtherdigital processing, decision making and decoding in unit 31.4. The datais provided to output port 31.5.

FIG. 32 shows Antenna Arrays (AA) in this multiple transmit and receivewith omni and/or directional antenna embodiment. This architecture withAdaptive Antenna Arrays (AAA) has the potential to increases the NLAspectral efficiency of FQPSK, FQAM and FGMSK systems to more than 30b/s/Hz by processing the transmit and/or receive antenna signals in adirectional mode. An interesting prior art reference is Winters [23],which contains several system applications of system applications ofconventional non-patented modulated systems. The potential advantage ofthe embodiment of FIG. 32 for FQAM and FGMSK systems as due to the factthat the FQPSK and FQAM type of systems are suitable for NLA power andRF spectral efficient robust performance operation with single, dual andmultiple antennas. Larger number of AA further increases the spectralefficiency and thus capacity of the systems of this invention inspecific geographic areas covered by the AAs and AAAs.

FIG. 33 shows a Pseudo-Error (PE) on-line, or in-service monitor genericblock diagram with on-line Probability of Error (Pe) monitor, diversitycontrol, adaptive equalization control and Non-Redundant Error Controlcircuit also designated as Feher's Non-Redundant Error Control (FN)detection and correction. Non-intrusive on-line, also known asin-service bit error or Bit Error Rate (BER) or Probability of Error(Pe) monitoring of received signals has numerous benefits for the userof digital communication systems. Prior art references including Feher's[1,2,3] and the references contained therein describe the operationalprinciples and some of the embodiments of the so-called “Pseudo-Error”on-line bit error detectors.” While pseudo-error detection by itself isknown from the prior art, pseudo-error monitoring of cross-correlatedand filtered BRA systems combined with pseudo-error monitor generatedcontrol signal generation for adaptive equalization and/or diversitycontrol techniques and embodiments are part of this new invention.

Pseudo-Error (PE) operation, adaptive equalization and diversity controlsignal generation by PE generated signals is described in relation toFIG. 33. On lead 33.1 the received RF signal is present. This signalcould be converted to a convenient IF and/or near the baseband frequencyrange and could be RF selective faded and/or corrupted by interference.In FIG. 33 an FQPSK received modulated signal is illustrated; othertypes of modulated signals could be also processed with the PE basedstructures of FIG. 33. In addition to signal distortion, exhibited interms of ISI, caused by RF selective, time-dispersive propagationchannels Interference 33.2 and Additive White Gaussian Noise (AWGN) 33.3is assumed to be corrupting this system by having interference and AWGNadded to the desired signal path in 33.4. Following the selective fadedsignal path and interference and noise, the power splitter 33.5 providesthe signal to I Channel Demodulator 33.7, 33.8 and 33.9 and Q channeldemodulator and to the Carrier Recovery (CR) unit 33.6. The regulardemodulated and regenerated data is present as “Data out to parallel toserial converter” in the I and Q channels. The PE circuit or“Pseudo-error on-line detector” has at its input stage an input port,designated as a buffer amplifier 33.10, followed by the 33.11 LPF(Low-Pass Filter in the “pseudo” path (lower case “p” in theabbreviation) and the regenerator or threshold detector 33.12. Unit33.13 is a clocked Exclusive-OR (ExOR) circuit which has the regulardemodulated “Data out to parallel to serial converter” as one of itsdata inputs “In1” and has the regenerated data from the pseudo dataregenerator 33.12 as its 2^(nd) data input “In2”. The clocked ExORcircuit (the clock input is not drawn to the ExOR—to simplify thedrawing) 33.13 provides binary pulses to the Pp(e) to P(e) converter33.14. Unit 33.14 is implemented by simple logic and counter circuitsand/or as part of a micro-processor and by software, hardware orfirmware.

Pseudo-Error (PE) implementations and their operating principles havebeen described in the literature, including references [1,2,3]. Unit33.14 provides a signal to the Non Redundant Error Control (NREC)circuit 33.19, which is used, for display of the actual estimated Pe orBER of the data stream. Implementations of PE detectors, described inthe aforementioned references, or variations of PE detectorimplementations used in this invention serve as Adaptive Equalizer (AE)control signal generators and Control Generators for Diversity Combiningand/or fast synchronization systems. In one of the structures of thisinvention 33.14 provides one or more signal lines to the AdaptiveEqualizer Controller 33.17 and in turn to the Adaptive Equalizer (AE)33.18. The AE provides to output ports 33.20 and 33.21 control signals.

In FIG. 34 an implementation block diagram of a Pseudo-Error (PE)controlled IF adaptive equalizer of this invention, designated as FeherEqualizer (“FE”) is shown. The elements, described in conjunction withFIG. 33, described previously are used in this structure of FIG. 34. Atthe input lead 34.1 the received RF or IF signal is provided to splitter34.2. The splitter output, designated as i(t) is fed to a signalcombiner 34.4 b. The lower branch of the splitter provides an input tosignal multiplier 34.3, followed by delay element D1 and the multipliedand delayed signal is fed to the second input of combiner 34.4 b. Thecombiner output is fed to an IF to baseband quadrature demodulator unit34.5. The multiplier unit 34.3 receives at its second input one or morecontrol signal (s), designated as c1(t) and generated by the PE monitorloop. The aforementioned multiplier 34.3 serves as a controlled signalattenuator, that is, it provides a time variable attenuation (or timevariable gain) in the lower branch of the “one tap” adaptive IFequalizer comprising splitter 34.2, combiner 34.4 b, multiplier (alsodesignated as mixer) 34.3, delay element D1 34.4 a and combiner unit34.4 b.

The Demodulator 34.5 of FIG. 34 has a structure such as the I and Qchannel demodulators shown in FIG. 33. The demodulated “eye” diagram(for definitions and terms such as eye diagram see one of the referenceslisted, including [1, 2 or 3]) through interface 34.6 provides the dataoutput via threshold regenerator and logic 34.7 to output port 34.8 andthe clock to 34.9, while the signals to the PE monitor are on leads34.10 and 34.11. Single or multiple lead signals on lead(s) 34.13 areprovided to LPF 34.12 for signal shaping and/or processing and forproviding the aforementioned c1(t) control signal to one of the inputsof the multiplier 34.3 of the adaptive equalization system. Theoperation of the adaptive Feher Equalizer (FE) shown in FIG. 34 operatesin a baseband to IF feedback loop in which the control signal isgenerated by a PE detector. The PE signal is a binary fully regeneratedsignal thus it is not the same type of signal as used to control thecoefficients of conventional adaptive taps of adaptive equalizers.

FIG. 35 is a diagram of a multiple delay switchable Adaptive Equalizer(AE) designated as Feher Rake “FR”. A Non-Redundant Error Control (NEC)detector, such as a Pseudo-Error (PE) circuit based NEC is used togenerate one or multiple control signals, designated as “Control SelectSignals”. Received signal on lead 35.1 is provided to multiple splitterports 35.2,35.6 and 35.10. The splitters provide signals to variablegain amplifiers A1 to An to provide signals to delay components D1 toDn. The respective components are: 35.2 to 35.14. The upper and lowerbranches of the split signals are re-combined in adders 35.3, 35.7 and35.11. The signal selection switch, Switch unit 35.14 selects one of theIF signals present at 35.3, 35.7 and 35.11 combiner outputs. The signalselection is controlled by the “Control Select Signals” generated byunit 35.21. The selected signal at the output port of switch 35.14 isprovided to 35.15 demodulator and PE monitor NEC detector. Output port35.16 provides signals to 35.19 Pseudo Error NEC detector additionalprocessing and in turn for signal shaping by a LPF 35.20 for providingsignals to the Control Select Generator 35.21. The FR is a combinationof an adaptive Feher Equalizer (FE) with selectable and switchable delaycomponents among several branches or “rake branches” of the receiverstructure. In this embodiment the receiver operates based on theprinciple that if the PE detector and/or alternate NEC circuit has arelatively high Pe on line state then one or more of the componentvalues of the delay elements will be switched out or in, that isselected, and similarly the gain values of the adaptive equalizer A1,A2, . . . , An will be changed, continuously or in discrete steps.

In FIG. 36 an implementation architecture for multiple Adaptive FeherEqualizers (FE) and Feher Rakes (FR) with one or more demodulators isillustrated. The received signal on lead 36.1 is provided to splitterport 36.2 and the splitter provides signals to units 36.3, 36.4 and36.5. These units are FE and FR based on the previously describedembodiments. These units provide signals to units 36.6,36.7 and 36.8 formultiple signal demodulation. Single or multiple PE monitor(s) 36.10provide control signals to select by selection switch 36.9 the bestsignal.

FIG. 37 shows a 2-branch diversity receiver with an adaptive equalizerand a single demodulator. In this embodiment FQPSK and/or FQPSK type ofsignals are first combined, in combiner unit 37.6 and are afterwardsAdaptively Equalized in unit 37.7. Receiver antennas 37.1 and 37.2provide signals to RF mixers/down-converters 37.3 and 37.5. Oscillatorand/or Frequency Synthesizer 37.4 provides Carrier Wave signals fordown-conversion. The IF down-converted signals are provided to combinerand/or switch unit 37.6, followed by adaptive equalizer 37.7, byoptional Linear or NLA amplifier 37.8 and by Demodulator 37.9. Theoutput demodulated data is available on port 37.10, the received/bitsynchronized clock on port 37.11 and additional timing, such as symboltiming or block sequence timing on 37.12. One or more of the units andelements of this invention, described in conjunction with previousfigures are used in the architecture of this FIG. 37. The PE and/orother NEC generated control signals provide for best possible combining,including “Equal Gain Combining”, “Weighted Gain Combining”, “SelectiveCombining” and other combining methods described in the prior artliterature. A fundamental difference is that in this invention theactual Pseudo-Error rate is used as the pre dominant control signalgenerator and the PE monitor selects and/or combines the bestperformance BER signals, while the prior art combiners select or combinebased on the received Carrier Power or received Carrier-to-Noise (C N)ratio. Combining or selecting signals, based on received C/N or receivedcarrier power (C) may lead to the selection of the inferior performancechannel or combing with the wrong ratio or wrong weight, and inparticular if the RF channel has serious frequency selective fades. Forexample the “main RF signal” designated RF main from antenna 37.1 has inone instant a much higher received carrier power than the carrier poweron RF diversity antenna 37.2. Based on prior art receivers, combinersand diversity selection criteria the “main” signal would be selected asit has a higher C and thus higher C/N. However, in a severe RF selectivefaded environment even though the C power of the main branch, antenna37.1 is much larger than that of the diversity antenna 37.2. In thisinstance the main antenna signal has much more RF frequency selectivefade thus much worse performance the prior art receiver wouldchoose/select the main branch with its poor and inferior performance,while the PE and/or other NEC based control signal disclosed in thisinvention would select the diversity signal with its superiorperformance.

Additional Description

Having now described numerous embodiments of the inventive structure andmethod in connection with particular figures or groups of figures, andhaving set forth some of the advantages provided by the inventivestructure and method, we now highlight some specific embodiments havingparticular combinations of features. It should be noted that theembodiments described heretofore, as well as those highlighted belowinclude optional elements or features that are not essential to theoperation of the invention.

A first embodiment (1) provides a bit rate agile communication systemcomprises a splitter receiving an input signal and splitting the inputsignal into a plurality of baseband signal streams; a baseband signalprocessing network receiving the plurality of baseband signal streamsand generating cross-correlated cascaded processed and filtered bit rateagile (BRA) in-phase and quadrature-phase baseband signals; and aquadrature modulator receiving and quadrature modulating thecross-correlated filtered in-phase and quadrature-phase baseband signalsto generate a quadrature modulated output signal.

A second embodiment (2) further requires of the bit rate agilecommunication system that the baseband signal processing networkincludes a cross-correlator and at least one bit rate agile cascadedmismatched (ACM) modulator filter.

A third embodiment (3) further requires of the bit rate agilecommunication system that it comprise: a demodulator structure having atleast one bit rate agile (BRA) cascaded mismatched (ACM) demodulationfilter which is mismatched (MM) to the cascaded processed and filteredmodulated signal, and operating to demodulate the bit rate agile signal.

A fourth embodiment (4) further requires of the bit rate agilecommunication system that the at least one processed and filteredbaseband signal is generated by a plurality of modulator filters, and atleast one bit rate agile (BRA) demodulator filter is used for signaldemodulation.

A fifth embodiment (5) further requires of the bit rate agilecommunication system that the plurality of modulator filters, and thedemodulator filter are connected in either serial, parallel, or acombination of serial and parallel topology.

A sixth embodiment (6) provides bit rate agile communication systemcomprising:

a baseband signal processing network receiving parallel baseband signalstreams and generating combined Time Constrained Signal (TCS) responseand Long Response (LR) filtered in-phase and quadrature-phase basebandsignals; and a quadrature modulator receiving and quadrature modulatingthe Time Constrained Signal (TCS) response and Long Response (LR)filtered in-phase and quadrature-phase baseband signals to generate aquadrature modulated bit rate agile output signal.

A seventh embodiment (7) further requires that the bit rate agile systemfurther comprise: a transmit amplifier receiving the quadraturemodulated output signal and generating an amplified transmit signal forcoupling to a transmission medium.

An eighth embodiment (8) further requires that the bit rate agile systemfurther comprising a demodulator receiving and demodulating the bit rateagile transmit signal.

A ninth embodiment (9) provides in a communication system, a method forgenerating bit rate agile signals comprising steps of: receiving aninput signal and converting the input signal into a plurality of signalstreams; processing the plurality of signal streams to generatecross-correlated signals having changeable amounts of filtering for bitrate agile in-phase and quadrature-phase baseband signals; andmodulating the cross-correlated filtered in-phase and quadrature-phasebaseband signals to generate a quadrature modulated bit rate agileoutput signal.

A tenth embodiment (10) provides in a signal transmission system, amethod for generating bit rate agile signals comprising steps of:receiving a plurality of signal streams; processing the plurality ofsignal streams to generate cascaded Time Constrained Signal (TCS)response and Long Response (LR) filtered in-phase and quadrature-phasebaseband signals; and modulating the Time Constrained Signal (TCS)response and Long Response (LR) filtered in-phase and quadrature-phasebaseband signals to generate a quadrature modulated bit rate agileoutput signal.

An eleventh embodiment (11) provides a Bit Rate Agile (BRA) structurecomprising a input port for receiving input data; a splitter having aninput coupled to the input port, and serving to split the input datainto baseband signal streams; a baseband signal processing network forreceiving the baseband signal streams and providing cross-correlated andfiltered Bit Rate Agile (BRA) in phase and quadrature phase basebandsignals; a Quadrature Modulator serving to quadrature modulate thecross-correlated filtered in phase and quadrature phase basebandsignals; an interface transmitter port to provide the quadraturemodulated signal to the transmission medium; an interface receiver portto provide connection of the cross-correlated filtered quadraturemodulated signal to the demodulator; and a demodulator structure toserve for Bit Rate Agile (BRA) signal demodulation having Bit Rate Agile(BRA) demodulation filters Mis-Matched (MM) to that of the modulatorfilters.

A twelfth embodiment (12) further requires of the Bit Rate Agile (BRA)structure structure that the processed in phase and quadrature phasebaseband signals have amplitudes such that their vector sum issubstantially constant and has reduced resultant quadrature modulatedenvelope fluctuations.

A thirteenth embodiment (13) further requires that the Bit Rate Agile(BRA) structure comprises means for selectively reducing the crosscorrelating factor down to zero.

A fourteenth embodiment (14) provides a Bit Rate Agile (BRA) structurecomprising a baseband signal processing circuit receiving one or morebaseband signal streams and providing cross-correlated and filtered BitRate Agile (BRA) in-phase and quadrature-phase baseband signals; aquadrature modulator serving to quadrature modulate the cross-correlatedfiltered in phase and quadrature phase baseband signals; a transmitamplifier to provide the quadrature modulated signal to the transmissionmedium; an interface receiver port to provide connection of thecross-correlated filtered quadrature modulated signal to thedemodulator; and a demodulator structure to serve for Bit Rate Agile(BRA) signal demodulation.

A fifteenth embodiment (15) provides a Bit Rate Agile (BRA) structurecomprising: a baseband signal processing network for receiving basebandsignal streams and providing cascaded Bit Rate Agile (BRA) TimeConstrained Signal (TCS) response and Long Response (LR) filtered inphase and quadrature phase baseband signals; a Quadrature Modulatorserving to quadrature modulate the cascaded Time Constrained Signal(TCS) response and Long Response (LR) filtered in phase and quadraturephase baseband signals; an interface transmitter port to provide thequadrature modulated signal to the transmission medium; an interfacereceiver port to provide connection of the filtered quadrature modulatedsignal to the demodulator; and a demodulator structure to serve forsignal demodulation having Bit Rate Agile (BRA) demodulation filtersMis-Matched (MM) to that of the modulator filters.

A sixteenth embodiment (16) provides a Bit Rate Agile (BRA) structurecomprising: a input port for receiving input data; a splitter having aninput coupled to the input port, and serving to split the input datainto baseband signal streams; a baseband signal processing network forreceiving the baseband signal streams and providing cascaded TimeConstrained Signal (TCS) response and Long Response (LR) filtered inphase and quadrature phase baseband signals; a Quadrature Modulatorserving to quadrature modulate the Time Constrained Signal (TCS)response and Long Response (LR) filtered in phase and quadrature phasebaseband signals; a transmit amplifier to provide the quadraturemodulated signal to the transmission medium; an interface receiver portto provide connection of the filtered quadrature modulated signal to thedemodulator; and a demodulator structure to serve for Bit Rate Agile(BRA) signal demodulation.

A seventeenth embodiment (17) provides a structure comprising: an inputport for receiving baseband signals; a baseband signal processingnetwork for receiving the baseband signals and providingcross-correlated bit rate agile cascaded mismatched (ACM) processed andfiltered in-phase and quadrature-phase baseband signals.

A eighteenth embodiment (18) provides a signal processing, modulation,transmission, signal reception and demodulation system, for Bit RateAgile (BRA), Modulation Demodulation (Modem) Format Selectable (MFS) andCode Selectable (CS) systems comprising: (a) means for input port forreceiving input data; (b) splitter means serving for BRA, MFS and CSsignal splitting, having an input coupled to the input port, and servingto split the input data into baseband signal streams; (c) means for BRA,MFS and CS baseband signal processing; (d) means for receiving thebaseband signal stream and providing for BRA, MFS and CS systemschangeable amounts of cross-correlation between Time Constrained Signal(TCS) response processors combined with TCS and Long Response (LR)processors; (e) means for cross-correlated processed in phase (I) andquadrature (Q) phase baseband signals for quadrature modulation to the Iand Q input ports of the Quadrature Modulator (QM); (f) means for aninterface unit to provide the quadrature modulated data to thetransmission medium; (g) means for a receiver interface unit forconnection of the received cross-correlated signal to the BRA and MFSdemodulator; (h) means for BRA, MFS and CS demodulation; (i) means forpost-demodulation Mis-Matched (MM) filtering of the BRA MFS and CSdemodulated signals in which the MM demodulator filters are mismatchedto that of the BRA and MFS filters.

A nineteenth embodiment (19) provides a cross-correlated signalprocessor comprising: (a) means for Bit Rate Agile (BRA), andModulation-Demodulation (Modem) Format Selectable (MFS) input port forreceiving input data; (b) means for providing BRA and MFS in-phase (I)and quadrature phase (Q) signals; (c) BRA and MFS means forcross-correlating a fraction of a symbol or one or more symbols of the Isignal with one or more symbols of the Q signal; (d) means forimplementing the BRA and MFS cross-correlated signals by analog activecircuits, analog passive circuits, by digital circuits or anycombination thereof; (e) means for switching in-out additional filtersin the I and/or Q channels; (f) means for Quadrature modulating the Iand Q signals; (g) means for Linear and/or Nonlinear amplification toprovide to the antenna; (h) a receiver port for connection of thereceived cross-correlated signal to the BRA and MFS demodulator; (i) aBRA and MFS quadrature demodulator; and (j) a Mis-Matched (MM) BRA andMFS demodulator filter set in which the demodulator filter set is MM tothat of the BRA and MFS filter set of the modulator.

A twentieth embodiment (20) provides a cross-correlated signal processorfor Bit Rate Agile (BRA) and Modulation-Demodulation (Modem) FormatSelectable (MFS) and Code Selectable (CS) means comprising: (a) meansfor providing in-phase and quadrature phase signals; (b) means forcross-correlating a fraction of a symbol or one or more than one symbolof the in-phase (I) signal with a fraction of a symbol or one or morethan one symbol of the quadrature-phase (Q) signal; (c) means forgenerating filtered cross-correlated I and Q signals; (d) means forimplementing the cross-correlated signals by analog active or passivecircuits, by digital circuits or combination thereof; (e) means forproviding a control circuit to select from a set of predeterminedcross-correlated signal elements filters and selectable waveforms in theI and/or Q channels; (f) means for Quadrature modulating the I and Qsignals; (g) means for Linear and/or Nonlinear amplification to provideto the antenna; (h) a receiver port for connection of the receivedcross-correlated signal to the BRA and MFS demodulator; (i) a BRA andMFS quadrature demodulator; (j) a Mis-Matched (MM) BRA and MFSdemodulator filter set in which the demodulator filter set is MM to thatof the BRA and MFS filter set of the modulator.

A twenty-first embodiment (21) provides a Cross-correlated signalprocessor for Bit Rate Agile (BRA) and Modulation-Demodulation (Modem)Format Selectable (MFS) and Code Selectable (CS) means comprising: (a)processing means for one or more input signals and providing in-phase(I) and quadrature phase (Q) signals; (b) means for cross-correlatingthe in-phase and quadrature shifted signals; (c) means for generatingin-phase and quadrature shifted output signals having amplitudes suchthat the vector sum of the output signals is approximately the same atvirtually all phase angles of each bit period for one set ofcross-correlation and filter parameters and the vector sum is notconstant for an other set of chosen filter parameters; (d) means forquadrature modulating the in-phase and quadrature output signals, toprovide a cross-correlated modulated output signal; (e) means forproviding a control circuit to select from a set of predeterminedcross-correlated signal elements filters and selectable waveforms in theI and/or Q channels; (f) means for Quadrature modulating the I and Qsignals; (g) means for Linear and/or Nonlinear amplification to provideto the antenna; (h) a receiver port for connection of the receivedcross-correlated signal to the BRA and MFS demodulator; (i) a BRA andMFS quadrature demodulator; and (j) a Mis-Matched (MM) BRA and MFSdemodulator filter set in which the demodulator filter set is MM to thatof the BRA and MFS filter set of the modulator.

A twenty-second embodiment (22) provides a cross-correlated signalprocessor comprising: (a) means for cross-correlating a fraction, or oneor more than one symbol synchronous and/or asynchronous time constrainedsignal (TCS) response and cascaded long response (LR) filtered signalsymbols of one or more input signals with signal symbols of a quadraturephase shifted signal of the in-phase signal, and providing in-phase (I)and quadrature phase (Q) shifted signals for Bit Rate Agile (BRA),cascaded mismatched (ACM) Modulation-Demodulation (Modem) FormatSelectable (MFS) and Code Selectable (CS) processing, according to thefollowing schedule: (i) when the in-phase channel signal is zero, thequadrature shifted signal is close to the maximum amplitude normalizedto one (1); (ii) when the in-phase channel signal is non-zero, themaximum magnitude of the quadrature shifted signal is reduced from 1(normalized) to A, where 0_A_(—)1; (iii) when the quadrature channelsignal is zero, the in-phase signal close to the maximum amplitude; (iv)when the quadrature channel signal is non-zero, the in-phase signal isreduced from 1 (normalized) to A, where 0_A_(—)1; (b) means forquadrature modulating the in-phase and quadrature output signals toprovide a cross-correlated modulated output signal; (c) controllingmeans and signal selection means for BRA rate, MFS and CS processorselection and selection for Linear and/or Non-Linearly Amplified (NLA)baseband and/or of Quadrature modulated signals; (d) coupling port meansto the transmission medium; (e) a receiver port for connection of thereceived cross-correlated signal to the BRA, MFS and CS demodulator; (f)a BRA, MFS and CS quadrature demodulator; and (g) a Mis-Matched (MM)demodulator filter set for BRA, MFS and CS in which the demodulatorfilter set is MM to that of the BRA, MFS and BRA filter set of themodulator.

A twenty-third embodiment (23) provides a structure for trellis codingand decoding, of extended memory Bit Rate Agile (BRA),Modulation-Demodulation (Modem) Format Selectable (MFS) and CodeSelectable (CS) input port for receiving input data comprising: atrellis encoder; a BRA, MFS and CS splitter having an input coupled tothe input port, and serving to split the input data into baseband signalstreams; a BRA, MFS and CS baseband signal processing network forreceiving the baseband signal streams and providing BRA, MFS and CS inphase (I) and quadrature (Q) phase baseband signals to the I and Q inputports of the transmitter; means for baseband signal processing forreceiving the baseband signal streams and providing for BRA, MFS and CSsystems changeable amounts of cross-correlation; means for selectivelyreducing the cross-correlating factor down to zero between TimeConstrained Signal (TCS) response processors combined with TCS and LongResponse (LR) processors; a receiver port for connection of the receivedcross-correlated signal to the BRA and MFS demodulator; a BRA and MFSquadrature demodulator; and a Mis-Matched (MM) BRA and MFS demodulatorfilter set in which the demodulator filter set is MM to that of the BRAand MFS filter set of the modulator.

A twenty-fourth embodiment (24) provides a cross correlated quadraturearchitecture signal processor for producing Bit Rate Agile (BRA),cross-correlated in phase and quadrature phase signal streams formodulation by a Quadrature Modulator and transmission and for signaldemodulation comprising: (a) means for receiving an input BRA signalselected from the group of binary, multi-level, and analog signals andcombinations thereof; (b) filtering means of the BRA input signal; (c)BRA signal shaping means for the filtered input signal; (d)amplification means for varying the modulation index of the BRA signal,the amplifier receiving the filtered input signal and providing anamplified input signal; (e) means for BRA signal splitting for receivingthe amplified input signal; (f) cross correlation means of BRA datastreams; and a BRA signal processor means having an in phase andquadrature phase channel each receiving one of the cross-correlated datastreams, each of the in phase and quadrature phase channel having afirst delay gain filter, means for generating BRA Cosine and BRA Sinevalues for the in phase and quadrature phase channel data stream; (g) aBRA wave shaper and a second BRA delay gain filter, such that the signalprocessor provides in phase and quadrature phase cross correlated datasignal processor; (h) means for quadrature modulation with a BRAmodulated signal adaptable for coherent or non-coherent demodulation ofthe quadrature BRA Frequency Modulated (FM) signal; (i) controllingmeans and signal selection means for BRA rate processor selection; (j)selection means for Linear and/or Non-Linearly Amplified (NLA) basebandand/or of modulated signals coupling port means to the transmissionmedium; (k) receiver port means for connection of one or more receivedcross-correlated signals to the BRA demodulator; (1) BRA demodulatormeans; and (m) Mis-Matched (MM) demodulator filtering means for BRA, MFSand CS demodulation in which the demodulator filter set is MM to that ofthe BRA, MFS and BRA filter set of the modulator.

A twenty-fifth embodiment (25) provides a signal processing, modulation,transmission, signal reception and demodulation system, designated asFeher's Gaussian Minimum Shift Keyed (GMSK) for Bit Rate Agile (BRA),Modulation Demodulation (Modem) Format Selectable (MFS) and CodeSelectable (CS) systems comprising: (a) input port for receiving inputdata; (b) Gaussian low-pass filter and presetable gain integrator forprocessing the input data and providing filtered input data; (c) asplitter having an input coupled to the input port, and serving to splitthe filtered input data into in phase (I) and quadrature phase (Q)channel cross coupled data streams such that the I and Q data streamsare proportional in gain and phase to the input data; (d) a signalprocessing network for receiving the I and Q channel data streams andproviding processed in phase and quadrature phase signals, the signalprocessing network including a signal processor for varying themodulation index for the signal processing network; (e) means forgenerating Cosine and Sine values for the I and Q channel BRA, MFS andCS data streams; (f) means for filtering by bit rate agile FIR or IIR orswitched filter and/or other post GMSK shaping filters the signals inthe I and Q channels such that the signal processor provides in phaseand quadrature phase cross correlated data signals for quadraturemodulation with a modulated signal suitable for amplification in linearand non-linear mode; (g) means for providing the amplified signal to thetransmission port; (h) a receiver port for connection of the receivedcross-correlated signal to the BRA and MFS demodulator; (i) a BRA andMFS quadrature demodulator; and (j) a Mis-Matched (MM) BRA and MFSdemodulator filter set in which the demodulator filter set is MM to thatof the BRA and MFS filter set of the modulator.

A twenty-sixth embodiment (26) provides a structure comprising: a inputport for receiving baseband signals; and a baseband signal processingnetwork for receiving the baseband signals and providingcross-correlated bit rate agile Peak Limited (PL) in-phase andquadrature-phase baseband signals.

A twenty-seventh embodiment (27) provides a structure for OrthogonalFrequency Division Multiplexed (OFDM) signals comprising: a input portfor receiving OFDM baseband signals; and a baseband signal processingnetwork for receiving the baseband signals and providingcross-correlated filtered in-phase and quadrature-phase basebandsignals.

A twenty-eighth embodiment (28) provides a structure comprising: a inputport for receiving Orthogonal Frequency Division Multiplexed (OFDM)baseband signals; and a baseband signal processing network for receivingthe OFDM signals and providing cross-correlated filtered in-phase andquadrature-phase baseband signals.

A twenty-ninth embodiment (29) provides a structure comprising: a inputport for receiving Orthogonal Frequency Division Multiplexed (OFDM)baseband signals; and a baseband signal processing network for receivingthe OFDM signals and providing cross-correlated Peak Limited (PL)in-phase and quadrature-phase baseband signals.

A thirtieth embodiment (30) provides a structure comprising: an inputport for receiving baseband signals; a baseband signal processingnetwork for receiving the baseband signals and providing more than twostate cross-correlated filtered in-phase and quadrature-phase basebandsignals; a Quadrature Modulator serving to quadrature modulate thecross-correlated filtered in-phase and quadrature-phase basebandsignals; and a transmit amplifier to provide the quadrature modulatedsignal to the transmission medium.

A thirty-first embodiment (31) provides a Bit Rate Agile (BRA) structurecomprising: a input port for receiving single or plurality of basebandbinary input signals; a baseband signal processing network for receivingthe baseband binary signals and providing combined Time ConstrainedSignal (TCS) response and Long Response (LR) filtered multi-levelin-phase and quadrature-phase baseband signals; and a QuadratureModulator serving to quadrature modulate the Time Constrained Signal(TCS) response and Long Response (LR) filtered in-phase andquadrature-phase baseband signals; a transmit amplifier to provide thequadrature modulated signal to the transmission medium; an interfacereceiver port to provide connection of the filtered quadrature modulatedsignal to the demodulator; and a demodulator structure to serve forsignal demodulation.

A thirty-second embodiment (32) provides a structure comprising: a inputport for receiving a plurality of baseband signals; a baseband signalprocessing network for receiving the plurality of baseband signals andproviding cross-correlated filtered in-phase and quadrature-phasebaseband signals to two or more quadrature modulators for quadraturemodulation; a set of two or more transmit amplifiers to amplify andprovide the quadrature modulated signals for RF combining; and acombiner device for RF combining of the quadrature modulated amplifiedsignals.

A thirty-third embodiment (33) provides a structure comprising: a inputport for receiving a plurality of baseband signals; a baseband signalprocessing network for receiving the plurality of baseband signals andproviding in-phase and quadrature-phase filtered baseband signals to twoor more quadrature modulators for quadrature modulation; and a set oftwo or more transmit amplifiers to amplify and couple the quadraturemodulated amplified signals to two or more antennas.

A thirty-fourth embodiment (34) provides a structure comprising: a inputport for receiving baseband signals; a baseband signal processingnetwork for receiving and splitting the signals and for providingcross-correlated filtered in-phase and quadrature-phase baseband signalsto two or more quadrature modulators for quadrature modulation; and aset of two or more transmit amplifiers to amplify and provide thequadrature modulated amplified RF signals to an antenna array.

A thirty-fifth embodiment (35) provides a structure comprising: a signalprocessing network for receiving and splitting signals and for providingcascaded Time Constrained Signal (TCS) response and Long Response (LR)filtered in-phase and quadrature-phase baseband signals to two or morequadrature modulators for quadrature modulation; a set of two or moretransmit amplifiers to amplify and provide the quadrature modulatedamplified RF signals for RF combining; and a combiner device for RFcombining of the quadrature modulated amplified signals.

A thirty-sixth embodiment (36) provides a structure comprising: aninterface receiver port to provide connection of received Bit Rate Agile(BRA) cross-correlated filtered quadrature modulated signal to thedemodulator; and a demodulator structure to serve for signaldemodulation of the signal.

A thirty-seventh embodiment (37) provides an adaptive equalizerstructure comprising: an interface receiver port to provide connectionof received modulated signal to the pre-demodulation adaptive equalizer;a pre-demodulation adaptive equalizer structure comprising splitter,multiplier and delay structure for generating a control signal andreceived modulated signal time delayed product in one branch of thesplitter and coupling the signal time delayed product in one branch ofthe splitter and the received modulating signal in the other branch ofthe splitter to a signal combiner; a signal combiner structure forcombining the delayed control signal and received modulated signalproduct; a demodulator structure for demodulating the combined delayedcontrol signal and received modulated signal product; and a controlsignal processor for generation of and connection of the control signalto the product multiplier circuit.

A thirty-eighth embodiment (38) provides an adaptive equalizer andswitchable delay structure comprising: an interface receiver port toprovide connection of received modulated signal to a plurality ofsplitters, amplifiers, delay elements and signal combiners for signalselection of the received modulated signal; a demodulator structure fordemodulating the selected received modulated signal; and a controlsignal processor for generation of the control signal.

The invention further provides methods and procedures performed by thestructures, devices, apparatus, and systems described herein before, aswell as other embodiments incorporating combinations and subcombinationsof the structures highlighted above and described herein.

All publications including patents, pending patents and reports listedor mentioned* in these publications and/or in this patent/invention areherein incorporated by reference to the same extent as if eachpublication or report, or patent or pending patent and/or referenceslisted in these publications, reports, patents or pending patents werespecifically and individually indicated to be incorporated by reference.The invention now being fully described, it will be apparent to one ofordinary skill in the art that many changes and modifications can bemade thereto without departing from the spirit or scope of the appendedclaims.

1. A system comprising: a first processor, transmit filter and firstmodulator for processing and filtering a signal into time divisionmultiplexed (TDM) Gaussian filtered signal and for modulating saidGaussian signal into Gaussian Minimum Shift Keying (GMSK) modulatedsignal; a second processor and second modulator for processing a signalinto Orthogonal Frequency Division Multiplexed (OFDM) signal and formodulating said OFDM signal into OFDM modulated signal; a firstamplifier and first transmitter for nonlinearly amplifying and forproviding amplified signal to a first transmitter for transmission ofthe nonlinearly amplified GMSK modulated signal; and a second amplifierand second transmitter for linearly amplifying and for providingamplified signal to a second transmitter for transmission of thelinearly amplified OFDM modulated signal.
 2. The system of claim 1wherein said system comprises a receiver and demodulator system forreceiving and demodulating a transmitted GMSK modulated signal a receivefilter and receive processor for filtering, processing and providingdemodulated cross-correlated in-phase and quadrature-phase filteredsignal, wherein said receive filter is mis-matched to said transmitfilter.
 3. The system of claim 1 wherein said time division multiplexed(TDM) Gaussian filtered signal comprises a cross-correlated in-phase andquadrature-phase filtered signal.
 4. The system of claim 1 wherein saidtime division multiplexed (TDM) Gaussian filtered signal comprises across-correlated in-phase and quadrature-phase processed TimeConstrained Signal (TCS) response waveform and Long Response (LR)filtered signal.
 5. The system of claim 1 wherein said system comprisesa receiver a demodulator and two antennas.
 6. The system of claim 1wherein said system comprises a receiver a demodulator and comprises adiversity receiver.
 7. The system of claim 1 wherein said firsttransmitter and second transmitter transmit in different radio frequency(RF) bands.
 8. The system of claim 1 wherein said GMSK modulated signalis used in a cellular network and said OFDM modulated signal is used ina wireless network, wherein said cellular and said wireless networks aredifferent networks.
 9. The system of claim 1 wherein said systemcomprises two antennas, two receivers and two demodulators for receivingand demodulating a GMSK modulated and a OFDM modulated signal.
 10. Thesystem of claim 1 wherein said GMSK signal is provided to a cellularnetwork and said OFDM signal is provided to a wireless network, whereinsaid cellular network and said wireless network are different networks.11. The system of claim 1 wherein said Orthogonal Frequency DivisionMultiplexed (OFDM) signal is a cross-correlated in-phase andquadrature-phase filtered signal.
 12. The system of claim 1 wherein saidGMSK signal is a cross-correlated in-phase and quadrature-phase filteredsignal provided to a cellular network and said OFDM signal is across-correlated in-phase and quadrature-phase filtered signal providedto a wireless network, wherein said cellular network and said wirelessnetwork different networks.
 13. The system of claim 1 wherein saidsystem comprises receiver and demodulator for receiving and demodulatinga GMSK modulated and a OFDM modulated signal and for providingcross-correlated in-phase and quadrature-phase filtered GMSK and OFDMdemodulated signals.
 14. The system of claim 1 wherein said systemcomprises receiver demodulator and receiver filter for receiving anddemodulating a GMSK modulated and a OFDM modulated signal for providingcross-correlated in-phase and quadrature-phase filtered GMSK demodulatedsignal, wherein said receiver filter is for filtering of receiveddemodulated GMSK signal and is mis-matched to said transmit filter. 15.A method for: processing and filtering a signal into time divisionmultiplexed (TDM) Gaussian filtered signal and for modulating saidGaussian signal into Gaussian Minimum Shift Keying (GMSK) modulatedsignal; processing a signal into Orthogonal Frequency DivisionMultiplexed (OFDM) signal and for modulating said OFDM signal into OFDMmodulated signal; nonlinearly amplifying GMSK modulated signal forproviding nonlinearly amplified signal for transmission; and linearlyamplifying OFDM modulated signal for providing linearly amplified signalfor transmission.
 16. The method of claim 15 wherein said GMSK signal isa cross-correlated in-phase and quadrature-phase filtered signalprovided to a cellular network and said OFDM signal is across-correlated in-phase and quadrature-phase filtered signal providedto a wireless network, wherein said cellular network and said wirelessnetwork are different networks.
 17. The method of claim 15 wherein saidmethod comprises the steps of receiving and demodulating a GMSKmodulated and a OFDM modulated signal for providing cross-correlatedin-phase and quadrature-phase filtered GMSK or cross-correlated in-phaseand quadrature-phase OFDM demodulated signals.
 18. The method of claim15 wherein said method comprises steps of receiving, demodulating andfiltering a GMSK modulated and a OFDM modulated signal for providing areceived demodulated filtered transmitted GMSK modulated signal forproviding a cross-correlated in-phase and quadrature-phase filteredsignal, wherein said received filtered GMSK signal is mis-matched tosaid transmit filtered GMSK signal.